All weather tactical strike system (AWTSS) and method of operation

ABSTRACT

An AWTSS is shown to be made up of an improved synthetic aperture radar (SAR) for generating radar maps with various degrees of resolution required for navigation of an aircraft and detection of ground targets in the presence of electronic counter-measures and clutter. The SAR consists, in effect, of four frequency-agile radars sharing quadrants of a single array antenna mounted within a radome on a &#34;four axis&#34; gimbal with a sidelobe cancelling subarray mounted at the phase center of each quadrant. Motion sensors are also mounted on the single array antenna to provide signals for compensating for vibration and stored compensating signals are used to compensate for radome-induced errors. In addition, a signal processor is shown which is selectively operable to generate radar maps of any one of a number of desired degrees of resolution, such processor being adapted to operate in the presence of clutter or jamming signals.

BACKGROUND OF THE INVENTION

This invention pertains generally to air-to-ground guided missilesystems and particularly to systems of such type wherein a syntheticaperture radar is used to provide a map of optical quality of theterrain underlying an aircraft so that the position of ground targetsmay be determined with precision and an air-to-ground missile may beaccurately guided to impact on any selected one of such ground targets.

It has been postulated for some years that, in the field of guidancesystems for air-to-ground guided missiles, advantage could very well betaken of the speed and precision with which radar maps may be generatedby processing echo signals to a synthetic aperture radar to provideguidance systems having "all-weather" capability.

Any satisfactory "All Weather Tactical Strike System" (hereinafterreferred to by the acronym "AWTSS") must, in addition to possessing aguidance capability for air-to-ground guided missiles, have a capabilityof functioning as a navigational aid for the aircraft carrying the AWTSSand as a target acquisition and tracking means. All of the requiredfunctions, further, must be carried out in a hostile environment,meaning when the aircraft is maneuvering violently to avoid interdictingfire and when electronic countermeasures are being taken by the enemy.The fact that satisfactory operation in a hostile environment is arequisite has, to date made it impossible to provide an AWTSS which maybe used with any reasonable confidence of success.

SUMMARY OF THE INVENTION

With the foregoing background of this invention in mind, it is a primaryobject of this invention to provide an AWTSS which is capable ofperforming, in a hostile environment, all of the requisite actions ofsuch a system.

A further object of this invention is to provide an AWTSS which uses asingle radar to carry out the desired functions.

The foregoing and other objects of this invention are generally attainedby providing, in a tactical aircraft carrying air-to-ground missiles, animproved pulse Doppler radar and signal processor, which radar andprocessor operate in conjunction with internal measuring devices alsocarried on the aircraft to operate as desired in any of the modes ofoperation required for an AWTSS. Specifically, the contemplated radarand signal processor are arranged to be substantially immune toelectronic countermeasures and to be operative even when the aircraftequipped with the contemplated AWTSS is violently maneuvered.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of this invention, reference is nowmade to the following description of the accompanying drawings, wherein:

FIG. 1 is a block diagram showing generally the interconnections betweenthe major components of the contemplated AWTSS;

FIGS. 2A and 2B, taken together, constitute a block diagram of theexciter shown in FIG. 1 to be used;

FIG. 3A is a sketch illustrating how the antenna elements of a planararray are arranged to provide the quadrants and segments herecontemplated;

FIGS. 3B and 3C are views showing details of construction of the planararray of FIG. 3A;

FIGS. 4A and 4B, taken together, constitute a block diagram of the radarsynchronizer shown in FIG. 1, somewhat simplified to show how anexemplary receiver channel is controlled;

FIGS. 5 and 5A are diagrams of an exemplary channel of the ratiofrequency receiver showing radome calibration;

FIG. 6 is a block diagram of an exemplary channel of the intermediatefrequency receiver shown in FIG. 1;

FIG. 7A is a sketch illustrating how movement of an aircraft carryingthe contemplated AWTSS imposes limits on the extent of the field whichmay be satisfactorily mapped;

FIG. 7B is a sketch illustrating how the antenna of FIG. 1 may berotated about an axis to enlarge the field which may be satisfactorilymapped;

FIG. 7C is a block diagram showing how various elements of thecontemplated AWTSS may be interconnected to compensate for movement ofthe aircraft during a dwell;

FIGS. 8A and 8B are sketches showing how the gimbal axes of the antennashown in FIG. 1 are arranged and an outline drawing of the gimbalassembly supporting the antenna of FIG. 1;

FIG. 9A is a block diagram of the radar synchronizer of FIG. 1;

FIGS. 9B through 9F are block diagrams of various portions of the radarsynchronizer shown in FIGS. 1 and 9A;

FIG. 10 is a sketch showing, in a general manner, the geometry of atactical situation when the contemplated AWTSS is used to aid in thenavigation of an aircraft over a terrain;

FIGS. 11 and 11A are flow charts showing how signals received from,respectively, stationary and moving targets are contemplated to beprocessed;

FIGS. 12A and 12B are sketches showing how the flight path of anaircraft and the pulse repetition interval are contemplated to becontrolled during the navigation mode;

FIG. 13 is a sketch showing how the received signal strength from movingtargets varies with spacing between phase centers in the antenna ofFIGS. 3A, 3B and 3C and;

FIG. 14 is a flow showing how a high resolution map is generated.

DESCRIPTION OF THE PREFERRED EMBODIMENT General (FIG. 1)

Referring now to FIG. 1, the herein contemplated synthetic aperturepulse Doppler radar system 10 is shown generally to include an antenna14, a power distribution network 16, a multichannel radio frequencyreceiver (sometimes hereinafter referred to simply as R.F. receiver 18)and an inertial measuring unit 24, all of which are mounted on afour-axis gimbal assembly 12 and are to be described hereinbelow.Suffice it to say here that the antenna 14 has dual polarization and isdivided into four separate quadrants and four segments, as shown in FIG.3A, with a subarray of sidelobe cancelling elements in each separatequadrant. Radio frequency (R.F.) power from a transmitter 22 is fed tothe antenna 14 via the power distribution network 16 which, in thereceive mode of operation, directs the signals received by each antennaquadrant to the R.F. receiver 18. It is noted here in passing that theoutput signals from the subarrays are multiplexed into a single receiverchannel in a manner which will be described in detail hereinbelow. TheR.F. receiver 18 is effective to heterodyne the received signals tosuitable intermediate frequency (I.F.) signals which are subsequentlypassed to an I.F. receiver 20 mounted off the four-axis gimbal assembly12.

The inertial measuring unit 24 (sometimes hereinafter referred to as thehead mounted IMU 24) comprises a pair of cross accelerometers, aboresight accelerometer located on the back center of the antenna andthree mutually orthogonal rate gyroes. In addition, a total of fivevibration-sensitive piezoelectric accelerometers (not shown) areprovided directly on the rear face of the antenna 14, one at thegeometric center of such face and one at the center of each antennaquadrant. The output signals from the piezoelectric accelerometers (notshown) are passed to the R.F. receiver 18 wherein they are processed ina manner to be described in detail hereinbelow ultimately tophase-compensate the received signals to eliminate vibration-inducedsidebands.

The output signals from the I.F. receiver 20, converted to digitalformat, are sent to a signal processor 26 wherein motion compensationand spectrum analysis are performed in accordance with a Fast FourierTransform (FFT) algorithm. The resulting data are passed to a digitaldata processor 30 which serves as the command and control computer forthe radar system 10 and generates, inter alia, angle pointing commandsfor the four-axis gimbal assembly 12 and timing commands for a radarsynchronizer 34. It is noted here in passing that the requisite trackingloops (not shown) to maintain frequency control and to control theantenna 14 are implemented within the digital data processor 30.Completing the radar system 10 are an exciter 32 (which generates thetransmitted waveforms as well as the local oscillator signals for theR.F. receiver 18 and the I.F. receiver 20), a utilization device 40(which may, for example, be a cathode ray display tube), an inertialplatform 38, and a boresight sensor assembly 36. The latter two unitsprovide phase rotation multipliers to the signal processor 26 which areused for motion compensation of the received signals.

Exciter (FIGS 2A, 2B)

The transmit, receive and ECCM waveforms required to support the radarsystem 10 (FIG. 1) are generated in the exciter 32. The coherenttransmitted signals are either CW waveforms, frequency modulated (FM)chirp or binary phase modulated waveforms, sometimes referred tohereinafter as a Burst A waveform. In addition, a noncoherent waveformreferred to hereinafter as a Burst B waveform or a "spoof" waveform maybe transmitted as a counter-measure against jammers. Finally, therequisite local oscillator signals for coherently processing Burst Awaveforms are generated in the exciter 32.

Each kind of Burst A waveforms contains signals which arefrequency-agile from pulse-to-pulse following an ordered set of fourfrequencies. Each Burst B waveform is made up of a total of one to fouruncoded, frequency-agile, but noncoherent, pulses which are interlacedbetween pulses of a Burst A waveform.

Referring now to FIGS. 2A and 2B, control signals from the radarsynchronizer 34 (FIG. 1) are shown applied to a digital interfacecontrol unit 10. Such control signals are applied to the digitalinterface control unit 101 in serial format, non-real time. Controlsignals applied to the digital interface control unit 101 are storeduntil required by the various subassemblies of the exciter 32. A DATASTROBE pulse, also provided by the synchronizer 34 (FIG. 1), is theneffective to cause control signals to be taken out of the digitalinterface control unit after having been converted in a known manner toanalog signals whenever required.

Control signals from the digital interface control unit 101 are appliedto a frequency-agile generator 103, a waveform generator 105 and secondlocal oscillator (designated L.O. 2 generator 107). The output signalsfrom the frequency-agile generator 103 and the waveform generator 105are provided to an upconverter 109 (shown in greater detail in FIG. 2B)wherein the RF INPUT signal for the transmitter 22 (FIG. 1), the firstlocal oscillator signal (L.O. 1) for the RF receiver 18 (FIG. 1) and thePILOT PULSE signal are developed.

A 100 MHz COHO signal (the coherent local oscillator signal) is passeddirectly from the synchronizer 34 (FIG. 1) to a local oscillator (L.O. 3generator 111) to "phase lock" a 100 MHz voltage-controlled oscillator(not shown) in a conventional manner to provide a low noise COHO signalhaving both good long-term and short-term stability. The narrow bandcharacteristics of the phase lock loop (not shown) eliminateinterference picked up in the cable (not shown) between the synchronizer34 and the exciter 32 which would otherwise appear as jitter on thegenerated waveforms. The low noise COHO signal is provided as areference signal to the waveform generator 105 and L.O. 2 generator 107and, after appropriate multiplication to 500 MHz, is applied to thefrequency-agile generator 103 and also to phase shifters 351U, 351L(FIG. 6).

The frequency-agile generator 103 is of conventional design, here beingmade up of a total of four voltage-controlled oscillators (not shown)arranged so that a total of four tunable bandwidths of from 1 to 1.6 GHzmay be achieved. The output frequency of each of the voltage-controlledoscillators is determined in a known manner by an associated phase-lockloop (PLL) with the 500 MHz COHO signal from the L.O. 3 generator 111 asthe reference signal. The control signals to the frequency-agilegenerator 103 from the digital interface control unit 101 are convertedto equivalent offset voltages for each of the voltage-controlledoscillators. The resulting signals out of the voltage-controlledoscillators are then selectively heterodyned in a conventional mannerwith a reference signal at C-band (such signal being generated withinthe frequency-agile generator 103 in a conventional manner). Theresulting C-band signals are filtered and amplified to provide a set offrequency-agile signals to the upconverter 109.

The waveform generator 105, which is also of conventional design,generates modulating waveforms which are impressed on the output signalsfrom the frequency-agile generator 103 to produce the input signals tothe transmitter 22 (FIG. 1). Here the waveform generator 105 provideseither a binary phase-coded wave-form, a FM chirp waveform or a CWwaveform. To produce the binary phase-coded waveform a first portion ofthe 100 MHz COHO input signal to the waveform generator 105 ismultiplied by a factor of 11 in a frequency multiplier to produce a 1.1GHz signal. Such signal is band-pass filtered and applied to a PIN diodemodulator wherein phase is switched between 0° and 180° in response tocontrol signals from the digital interface control unit 101. The outputsignals from the PIN diode modulator are passed, via a single-poledouble-throw switch and an amplifier, as an input signal to theupconverter 109. The single-pole double-throw switch, in response tocontrol signals supplied by the digital interface control unit 101, iseffective to select the waveform applied to the upconverter 109. It willbe appreciated by those of skill in the art that a CW waveform forapplication to the upconverter 109 may be derived from the output of thewaveform generator 105 by not actuating the pin diode modulator. It willalso be appreciated that, as is done here for reasons to be explained, aCW waveform may be applied to the upconverter 109 by tapping the outputof the bandpass filter.

In order to produce an FM chirp waveform, a portion of the 100 MHz COHOinput signal to the waveform generator 105 is first multiplied by afactor of 72 in a frequency multiplier to produce a signal at 7.2 GHz. Avoltage-controlled oscillator is operated periodically to sweep thefrequency of its output signal from 8.3 to 8.9 GHz which is mixed with asignal at 7.2 GHZ to produce, inter alia, a signal (a chirp waveform)which periodically varies from 1.1 to 1.7 GHz. After appropriatefiltering, only the latter chirp waveform is passed through thesingle-pole double-throw switch to the upconverter 109.

At the upconverter 109 the 8.3 to 8.9 GHz output signal from thefrequency-agile frequency generator 103 is applied as an input signal toeach one of a pair of mixers 113 and 115. The second input signal tomixer 113 is a CW signal at 1.1 GHz from the waveform generator 105,such CW signal being taken from the output of the bandpass filter (notshown) in the waveform generator 105. The output signals from the mixer113 are passed, via a single-pole single throw switch 117, to a low passfilter 119. That filter allows only the lower sideband signal from themixer 113 to be passed so that a set of first local oscillator (L.O.)signals with frequencies 7.2 to 7.8 GHz appear at the output of the lowpass filter 119 corresponding with the signals output of thefrequency-agile frequency generator 103. The first L.O. signals arepassed, via an amplifier 121 and an isolator 123, to the R.F. receiver18 (FIG. 1).

The second input signal to the mixer 115 is the coded signal from thewaveform generator 105 (a CW, phase-coded or chirp waveform). The outputsignals from the mixer 115 are passed, via a single-pole single-throwswitch 125, to a high pass filter 127 which is effective to pass onlythe upper sideband signals from the mixer 115. Such filtered signals arepassed, via directional couplers 133, 139, an amplifier 129 and anisolator 131, to the RF transmitter 22 (FIG. 1). It will now beappreciated that the single-pole single-throw switches 117, 125 areactuated (by a control signal from the digital interface control unit101) so that the drive signal to the transmitter 22 (FIG. 1) is "off"when the RF receiver 18 (FIG. 1) is "on".

The directional coupler 133 and a single-pole single-throw switch 137are provided to allow Burst "B" waveforms from a Burst "B" V.C.O. 135 tobe passed to the transmitter 22 (FIG. 1). The Burst "B" generator 135incorporates a voltage-controlled oscillator whose frequency differsfrom any passing through the high pass filter 127. The actuation of thesingle-pole single-throw switch 137 is determined by control signalsprovided by the digital interface control unit 101.

The second directional coupler 139 is positioned to couple a portion ofthe signal to the transmitter 22 (FIG. 1) to a pilot pulse generator141. The latter, which is here of conventional design, is provided togenerate a simulated target return signal for the purpose of calibratingthe radar system 10 (FIG. 1). Thus, the pilot pulse generator 141 isshown to receive from the digital interface control unit 101, GAINFREQUENCY and RANGE commands. The GAIN command controls the gain of anamplifier (not shown) to provide simulated target return signals withdifferent amplitudes. The FREQUENCY command controls avoltage-controlled oscillator (not shown) whose output signal is mixedwith the coupled input signal in order to provide simulated targetreturn signals with different Doppler frequencies. The RANGE commandcontrols a variable delay line (also not shown) which is used tosimulate target return signals from targets at different ranges.

The L.O. 2 generator 107 is also of conventional design. This unitdevelops any of the requisite L.O. signals to down-convert receivedsignals in the I.F. receiver 20 (FIG. 1) to a second I.F. frequency. Tothis end, the 100 MHz COHO signal from the L.O. 3 generator 111 ismultiplied by a factor of 66 in a frequency multiplier to produce a 6.6GHz signal which is applied as a reference signal to a mixer (notshown). The second input signal to the latter is obtained from avoltage-controlled oscillator which is tunable from 8.3 to 8.9 GHz. Thecontrol signals to control that oscillator (as well as a ramp signalrequired to "dechirp" a received chirp waveform) are obtained from thedigital interface control unit 101. The signal from the mixer is passedthrough a 1.7 to 2.3 GHz low pass filter to reject the upper sidebandsignal from the mixer, amplified and then passed (via and isolator) to amixer 315 (FIG. 6) in the I.F. receiver 20 (FIG. 1).

TRANSMITTER (FIG. 1)

The transmitter 22 (FIG. 1) is of conventional design and comprises aliquid-cooled traveling wave tube (TWT) amplifier. A variable attenuator(not shown) and a drive amplifier (also not shown), both of which arecontrolled by the radar synchronizer 34 (FIG. 1), are provided on theinput to the TWT to provide a power management capability as well as toensure that the power from the TWT remains constant with frequency andthe TWT (not shown) does not saturate. The output power from the TWT ispassed via an isolator (not shown) and a harmonic filter (also notshown) to the power distribution network 16 (FIG. 1).

DUAl POLARIZATION ANTENNA (FIGS. 3A, 3B, 3C)

Referring now to FIGS. 3A to 3C, the dual polarization antenna 14comprises an array of 560 dual-polarized, stripline fed annular slotradiating elements 140 (sometimes hereinafter referred to simply aselements 140), which are capable of simultaneously supporting orthogonallinear polarizations. That is to say, in the transmit mode the elements140 radiate circularly polarized signals of either sense, while in thereceive mode the received circularly polarized signals are resolved intoorthogonal linearly polarized signals. The antenna aperture issubdivided into four identical quadrants 14A, 14B, 14C and 14D, eachcontaining 100 elements, and four identical segments 14AS, 14BS, 14CSand 14DS, each containing 40 elements. The center four elements of eachof the quadrants 14A, 14B, 14C and 14D are combined to form a sidelobecancelling subarray (not numbered) for their respective quadrants. Itshould be noted here that while the sidelobe cancelling subarrays (notnumbered) are utilized primarily in the receive mode, provision is madeto allow the former to be utilized in the transmit mode. It should alsobe noted here that the antenna aperture (not numbered) is divided intosquare quadrants rather than serrated quadrants even through the latterdesign would produce low quadrant sidelobes in the 0°, 45° and 90°planes of the antenna 14. If the quadrants 14A, 14B, 14C and 14D wereserrated the grating lobes would be randomly distributed to either sideof the principal and diagonal planes of the antenna 14. The squarequadrant configuration was here chosen because, in such a configuration,the sidelobes are known to lie along the diagonal planes so thatcompensation for such sidelobes may be effected.

The dual polarization antenna 14 and its requisite low and medium powerfeed networks (not shown) are contained entirely in two striplinepackages 143, 145, respectively. The elements 140 are annular slots, onefree-space wavelength in circumference, with an inter-element spacing of0.707 inches by 0.707 inches on a 45° grid. The elements 140 are etchedin the upper ground plane 147 of the first stripline package 143 whichalso contains the low power feed network. Mode suppression pins 149 areprovided around each of the radiating elements 140.

The low power feed network (not shown) comprises a plurality of reactivetype, binary 3 dB power dividers of conventional design, arranged toprovide each radiating element 140 with a pair (referred to as thevertically and horizontally polarized feed signals) of linearlypolarized feed signals in space and phase quadrature. Each of thereactive type power dividers feeds a total of eight radiating elements140 with the power for each such divider being derived from a mediumpower feed network contained in the second stripline package 145. Therequisite interconnections between the low and medium power feed network(not shown) are provided by means of feedthrough pins 151. Dielectricplugs 153 are inserted over the feedthrough pins 151 and metallic plates155 are provided over the dielectric plugs 153. Mode suppression pins157 are provided around each of the feedthrough pins 151. Coaxialconnectors 159 are attached in any convenient manner, as by means ofscrews (not numbered), to the stripline package 143 containing the lowpower feed network (not shown). Coaxial connectors 161 are similarlyattached to stripline package 145 containing the medium power feednetwork (also not shown). It should be noted here that the latternetwork provides a total of sixteen output signals (two vertically andtwo horizontally polarized signals from each one of the four antennaquadrants 14A, 14B, 14C and 14D) which are passed, via coaxial cables(not shown), to the power distribution network 16 (FIG. 1). A total of16 outputs, corresponding to two vertically and two horizontallypolarized signals from the sidelobe cancelling elements in each of thefour quadrants of the antenna 14, are provided by the low power feednetwork (not shown) and are passed, via coaxial connectors 159, andcoaxial cables (not shown), to the R.F. receiver 18 (FIG. 1).

The stripline packages 143, 145 are bonded together to form a compositeassembly mounted to an aluminum honeycomb support structure 163. Suchstructure then renders the stripline packages 143, 145 impervious to theeffects of vibration and, at the same time, provides a support memberfor the power distribution network 16 (FIG. 1) and the R.F. receiver 18(FIG. 1).

A small monopole antenna (not shown) is disposed at the geometric centerof the antenna 14 and is excited by the PILOT PULSE signals provided bythe power distribution network 16 when the antenna 14 is to becalibrated.

POWER DISTRIBUTION NETWORK (FIGS. 4A and 4B)

Before proceeding with a detailed description of the power distributionnetwork 16 it is noted that, for the sake of simplicity, only oneportion of that network (the portion required to feed a single antennaquadrant, here quadrant 14A), is shown in detail. Similarly, only thesidelobe canecelling networks for antenna quadrant 14A are shown indetail. It should be appreciated that the sidelobe cancelling networksfor the remaining quadrants, as well as the requisite hardware withinthe power distribution network 16 for feeding the remaining antennaquadrants, are identical to the portion shown and to be described.Finally, before proceeding, it should be noted that antenna segments14CS and 14DS are used only on transmit and, therefore, there are someminor differences (to be described) between the segment feeds.

Referring now to FIGS. 4A, 4B, power from the transmitter 22 (FIG. 1)axis waveguide rotary joint (not shown). A waveguide coupler 231 couplesoff a portion of the input power to the power distribution network 16for antenna segment illumination. It is noted here in passing that thehigh power feed portion (not numbered) of the power distribution network16 is implemented using half-height aluminum waveguide networks. Also,because of the high power airborne environment the waveguide ispressurized. Following the coupler 231 in the antenna segment feednetwork (not numbered) are a series of power dividers 233A, 233B and233C which are here reactive H-plane tees and which serve to divide theinput power and deliver it to a series of single-pole single-throwwaveguide switches 235A, 235B, 235C and 235D. The latter are controlledby control signals from the radar synchronizer 34 (FIG. 1). It should benoted here that, for the sake of drawing clarity, all control signalsfrom the radar synchronizer 35 will be represented simply as smallarrows on the units being controlled and the requisite digital-to-analog(D/A) converters will not be shown.

The switches 235A . . . 235D provide the radar system 10 (FIG. 1) with apower management and beam shaping feature which is required in somemodes (to be described in detail hereinafter) of the system. Suffice itto say here that by controlling the position of switches 237A . . . 237Din the antenna quadrant feed network (not numbered) and switches 235A .. . 235D in the antenna segment feed network (also not numbered), it ispossible to transmit through any portion, or portions, of the antenna 14(FIG. 3A) required for a particular mode. The output signals fromswitches 235C and 235D are passed, respectively, to waveguide powerdividers 239C, 239D which are also reactive H-plane tees. The outputarms (not numbered) of the latter are terminated in coaxial connectors(not shown) and the interconnections between antenna segments 14CS, 14DSand power dividers 239C, 239D are via semi-rigid coaxial cables (notshown).

As mentioned hereinabove, in certain system operating modes, it isdesirable to transmit through only one or two of the antenna segments14AS . . . 14DS. In these modes the waveguide switches 237A . . . 237Din the quadrant feed network (not numbered) as well as the requisiteswitches in the segment feed network (also not numbered) aresynchronously switched to their open position so that the power incidentthereon may be reflected back to an isolator (not shown) within thetransmitter 22 (FIG. 1) where adequate cooling is provided. The switchesare synchronously switched to prevent the dissipation of large amountsof power in the loads (not numbered) provided on the power dividers 233A. . . 233C and 241A . . . 241C. The quadrant switching function is notprovided by a single switch preceding power dividers 241A . . . 241Cbecause a single switch having the requisite power handling capabilitycould not be packaged within the allocated volume.

The output signals from switches 235A, 235B are split in waveguide powerdividers 239A, 239B and passed via waveguide, three-port circulators243A . . . 243D as input signals to antenna segments 14AS and 14BS. Theinterconnections between antenna segments 14AS, 14BS and the three-portwaveguide circulators 243A . . . 243D are made by semi-rigid coaxialcables (not shown). It should be noted here that all the segment feednetworks (not numbered) are made to be of equal electrical path lengthand that two inputs, corresponding to two orthogonal linearpolarizations, are provided to each of the antenna segments 14AS, 14BS,14CS and 14DS.

The three port waveguide circulators 243A . . . 243D serve as duplexersto pass, in the receive mode, the vertically and horizontally polarizedsignals from antenna segments 14AS, 14BS to switch networks 245V, 245H(where the letters V and H refer to the vertical and horizontalcomponents of polarization as resolved by the element feed networks (notshown)). It should be noted here that each of the switch networks 245V,245H includes a 180° hybrid (not shown) and, therefore, each suchnetwork may, in response to a control signal provided by the radarsynchronizer 34 (FIG. 1) produce an output signal corresponding to: (a)the input signal received from antenna segment 14AS; (b) the inputsignal received from antenna segment 14BS; or (c), the differencebetween the input signals received from antenna segments 14AS and and14BS. The signals out of switch networks 245V, 245H are passed as inputsignals to switching networks 247V, 247H, respectively.

The quadrant feed network (not numbered) includes a series of high-powerwaveguide power dividers 241A . . . 241C which are here "magic-tees"whose E-ports are terminated by loads (not numbered) and which areeffective in splitting the input power into four equal phase, equalamplitude output signals which are passed to the waveguide switches 237A. . . 237D. As mentioned hereinabove, only the hardware required forfeeding antenna quadrant 14A will be described in detail. Thus, theoutput signal from switch 237A is passed to a waveguide "magic-tee"power divider 249 wherein it is split into a pair of equal amplitude andequal phase output signals which are passed to a pair of three portwaveguide circulators 251V, 251H. The remaining two ports on each of thelatter are terminated in waveguide-to-coax adaptors (not shown). Thus,the output signals from the circulators 251V, 251H are passed to a pairof stripline power dividers 253V, 253H via coaxial cables (not shown).The latter dividers provide two pairs of input signals to the elementfeed networks (not shown) in antenna quadrant 14A (FIG. 3A).

It is noted here in passing that the transmission line networks (notnumbered) between power dividers 253V, 253H and circulators 251V, 251Hcontain two pairs of directional couplers 255V, 255H and 257V, 257H. Thedirectional couplers 255V, 255H are used in the receive mode ofoperation to inject pilot pulse signals during calibration of the radarsystem 10 (FIG. 1). That is to say, since the herein-contemplated radarsystem 10 (FIG. 1) utilizes a so-called "pipeline" receiver wherein thesignals received by each of the antenna quadrants 14A . . . 14D (FIG.3A) are brought down to the signal processor 26 (FIG. 1) in separatechannels rather than being combined in a monopulse arithmetic networkinto sum (Σ) and difference (Δ) channels, each of the separate channelsmust be calibrated to ensure that phase and amplitude tracking ismaintained within very precise tolerences. Such calibration is hereprovided through the use of pilot pulses. The pilot pulses are obtainedfrom the exciter 32 (FIG. 1) and are split in a power splitting network261. The pilot pulses then are of equal amplitude and phase and,therefore, any detected differences between the channels may then becompensated for within the signal processor 26 (FIG. 1).

It should be noted here that one of the output signals from the powersplitting network 261 is passed, via a circulator 262, to the monopoleantenna at the center of the antenna 14 (FIG. 1). As mentioned brieflyhereinabove, that monopole antenna (not shown) is excited by the appliedpilot pulse signal to calibrate the antenna 14 (FIG. 1). In order tominimize the effect of the antenna radome (not shown) on the antenna 14(FIG. 1) during the calibration procedure, the latter is alwaysgimballed to the identical position for purposes of calibration. In thismanner the antenna 14 (FIG. 1) will always receive the identicalreflections from the radome (not shown) and the effect of the lattermay, therefore, be neglected.

As mentioned hereinbefore, the center four elements of each of antennaquadrants 14A . . . 14D (FIG. 3A) are combined to form a sidelobecancelling subarray for their respective quadrants. These sidelobecancelling subarrays are utilized only in the receive mode and theiroperation in each of the quadrants is identical with the exception of anadded feature (to be described) provided between the subarrays inantenna quadrants 14A and 14B. As also mentioned hereinabove, only thesidelobe cancelling networks (not numbered) for antenna quadrant 14Awill be described in detail. Thus, the vertically and horizontallypolarized signals from the sidelobe cancelling elements of quadrant 14Aare combined in monopulse arithmetic networks 259V, 259H, respectively.The monopulse sum (Σ) and difference (Δ) signals then formed are passedto switches 261V, 261H which are responsive to control signals providedby the radar synchronizer 34 (FIG. 1) to gate either the sum or thedifference signals for further processing. The signals out of theswitches 261V, 261H are passed through a pair of directional couplers263V, 263H which couple off a portion of the signals from the switches261V, 261H. Such coupled signals are passed, via the switching networks247V, 247H, through the auxiliary channel (not shown) of the R. F.receiver 18 (FIG. 1) to the signal processor 26 (FIG. 1). The Σ and Δsignals from the switches 261V, 261H are also passed to sidelobecancelling circuits (not numbered) comprising variable attenuators 265V,265H and phase shifters 267V, 267H, all of which are of conventionaldesign and are controlled by control signals supplied by the radarsynchronizer 34 (FIG. 1). The output signals from the sidelobecancelling circuits (not numbered) are injected, via the directionalcouplers 257V, 257H, into the quadrant receiver channels (not numbered).

It will now be appreciated by those of skill in the art that sidelobecancellers are utilized to reduce the level of jamming power enteringthe main channel sidelobes of radar systems. That is to say, themagnitude and phase of the jamming signals received by the sidelobecancelling circuits (not numbered) are modified by the variableattenuators 265V, 265H and the phase shifters 267V, 267H in response tothe applied control signals so that, when such modified signals areinjected into the quadrant receiver channels (not numbered) they are ofequal magnitude but 180° out-of-phase with the jamming signals receivedby quadrant 14A. The requisite control signals are developed within thedigital data processor 30 (FIG. 1) from analysis of the Σ channelsignals received by the R. F. receiver 18 (FIG. 1) from the directionalcouplers 277V, 277H.

Digressing now for a moment, it will also be appreciated by those ofskill in the art that, in general, the sidelobe antennas also receivereturns from the target being tracked and inject such target returnsignals, together with the jammer cancellation signals, into the channelbeing nulled. Since the phase of the injected target return signals isrotated by the sidelobe canceller phase angle, a bias error willdevelop. However, because the herein-contemplated radar system 10(FIG. 1) is continuously self-calibrated by pilot pulses to the signalprocessor 26 (FIG. 1), and because the commanded amplitude and phase ofthe sidelobe cancelling circuits (not numbered) are known a priori bythe digital data processor 30 (FIG. 1), the bias error due to sidelobecancelling may be neutralized. Thus, the digital commands from thesignal processor 26 (FIG. 1) to the sidelobe cancelling circuits (notnumbered) are also sent to the signal processor 26 (FIG. 1) forcompensation purposes, thereby ensuring that the bias error on targetsignals is corrected.

It should be noted here that the herein-contemplated sidelobe cancellingcircuits (not numbered) provide an advantage in terms of wideinstantaneous bandwidth. That is to say, conventional sidelobecancellers incorporate cancelling antennas which are physicallydisplaced from the aperture to be cancelled. Such displacement resultsin a time delay which induces a phase dispersion across a finitebandwidth, thereby limiting the degree of nulling attainable in suchcancelling systems. Because the sidelobe cancelling subarrays are herecentered in the antenna aperture and because the cancelling is realizedat microwave frequencies rather than at I. F. frequencies as in standardsidelobe cancellers, a unique solution to the problem of phasedispersion is provided. Further, not only are the centered sidelobecancelling subarrays positioned to reduce phase dispersion, each needsto operate on a smaller aperture (a quadrant only as opposed to theentire antenna aperture). The single antenna quadrants 14A . . . 14D(FIG. 3A) have twice the sidelobe frequency stability as that of thefull aperture sum beam and therefore amplitude dispersal effects areminimized. That is, the null produced on a small quadrant by a centeredsidelobe canceller is twice as wide in angle as that normally obtainedfor full aperture cancelling and, therefore, cancelling over a greaterarea is realized. This feature is advantageous for handling multiplespatially displaced jammers and/or multipath effects.

The herein-contemplated radar system 10 (FIG. 1) also includes asidelobe monitor (to be described) to prevent false target repeater typejammers from overloading the target tracking capability of the system byinjecting false target returns through the antenna sidelobes, therebypreventing true target identification and tracking. Briefly, when thesidelobe monitor (not numbered) is utilized, two simultaneous maps of agiven area are generated within the digital data processor 30 (FIG. 1),a first one of such maps being a conventional map generated from thedata received by the four antenna quadrants 14A through 14D (FIG. 3A),and the second being generated from the data received through thesidelobe monitors (not numbered). The two maps are compared within thedigital data processor 30 (FIG. 1) and any potential targets whoseamplitude is greater in the sidelobe monitor generated map than in theconventional map are rejected as false targets. A moment's thought hereshould make it apparent that, as the false target repeater type jammersenter the conventional map via the antenna sidelobes, their amplitude inthat map vis-a-vis their amplitude in the sidelobes monitor generatedmap will be decreased by the ratio of the gain of the sidelobe monitorantenna pattern to the gain of the antenna sidelobes. That is to say,the gain of the sidelobe monitor pattern is designed to be greater thanthat of the antenna sidelobes in the region of the antenna sidelobes.

The output signals from the sidelobe monitor (not numbered) are passedthrough the auxiliary channel of the R. F. receiver 18 (FIG. 1).Directional couplers 269V, 269H are included in the transmission linesections (not numbered) carrying the signals from the sidelobecancelling subarrays to the switching networks 247V, 247H. Preceding thedirectional coupler 269V are a variable attenuator 271V, a switch 273SVand a phase shifter 273V, while the directional coupler 269H is precededby a variable attenuator 271H, a switch 273SH and a phase shifter 273H.The input signals to the variable attenuators 271V, 271H are obtainedfrom directional couplers (not shown) disposed, respectively, in thevertically and horizontally polarized sidelobe cancelling channels (alsonot shown) carrying the sidelobe cancelling signals from antennaquadrant 14B to the switching networks 247V, 247H. The variableattenuators 271V, 271H, the switches 273SV, 273SH and the phase shifters273V, 273H are all controlled by control signals supplied by the digitaldata processor 30 (FIG. 1). It should now be appreciated by those ofskill in the art that the variable attenuators 271V, 271H and the phaseshifters 273V, 273H provide the sidelobe monitor (not numbered) with anull forming capability which is useful in cancelling noise jammerswhich might otherwise render the latter ineffective.

Digressing now for a moment, it will be appreciated by those of skill inthe art that synthetic aperture radar maps are particularly vulnerableto dedicated tracking standoff spot jammers. That is to say, since mostsynthetic aperture radars employ waveforms having necessarily longcoherent dwell times to obtain azimuth angle resolution, they normallycannot shift frequency during a dwell in order to avoid spot jammernoise. Thus, the herein-contemplated synthetic aperture radar system 10(FIG. 1) employs a multiple frequency waveform comprising four coherentfrequencies, as was mentioned hereinabove. Although the use of multiplefrequency waveforms can defeat a single spot jammer, additional immunityis required to counter the threat of multiple spot jammers.

The sidelobe cancelling circuits (not numbered) will step in frequencywith the multiple frequency waveform and will form time-multiplexedspatial nulls in each resolved carrier frequency band. Thus, thesidelobe cancelling circuits (not numbered) will self-adapt andneutralize jamming in each carrier band; however, they will only besuccessful when the carrier channel is not being jammed by more than onejammer. Thus, two triple frequency jammers can double cover a maximum ofthree frequencies and essentially disable the sidelobe cancelling inthese channels, but the fourth carrier channel would still be clear.

Additional immunity from spot jammers is provided in theherein-contemplated system through the transmission of four spooffrequencies. As explained hereinabove, the exciter 32 (FIG. 1) maygenerate interlaced bursts of four frequency agile coherent pulses andfour frequency agile noncoherent pulse which are used as spoof frequencywaveforms. The four noncoherent spoofing frequency waveforms are narrowpulses to conserve on power and to fit within the coherent pulserepetition interval. The spoof waveforms are radiated through antennasegments 14AS and 14BS so that the spoof peak power will appear largerthan that of the coherent synthetic aperture radar waveforms. The spooftransmissions precede a synthetic aperture radar dwell time in order tocapture a multiple frequency spot jammer throughout the dwell time.

As was explained above, the switching networks 247V, 247H are effectiveto gate selected ones of the signals incident thereon to the fifth orauxiliary channel of the R. F. receiver 18 (FIG. 1). Directionalcouplers 277V, 277H are provided in the output arms (not numbered) ofthe former for the purpose of injecting pilot pulse signals into theauxiliary channel of the R. F. receiver 18 (FIG. 1).

Finally, before proceeding it should be noted that a third waveguidedirectional coupler 279 is provided in the input channel of the powerdistribution network 16. The former couples off a portion of the inputpower to a power splitting network 281 which divides said portion intofive equal amplitude and equal phase output signals, referred to here asREPLICA signals. The REPLICA signals are provided to compensate for theeffects of vibration on the waveguide components in the powerdistribution network 16 which otherwise would be manifested in the formof false targets or paired echoes appearing in the Doppler band on eachside of the return signals. That is to say, vibrations of the R. F.components impress an amplitude and phase modulation on a chirp signal,thereby producing deviations from ideal quadratic phase characteristicswhich distinguish a chirp waveform. A second problem, existing with anytype of transmitted waveform, is that the phases of successive pulsesvary randomly. Because each one of the REPLICA signals corresponds witha transmitted signal, successively derived REPLICA signals may bedemodulated, converted to digital numbers and processed to form (foreach receiver channel) a running "nominal" complex conjugate of thetransmitted pulses. When a received signal is multiplied by the nominalcomplex conjugate existing as the received signal is being processed inthe I. F. receiver 20 (FIG. 1), the resulting signal is compensated fordeviations in phase or amplitude of the corresponding transmittedsignal.

R. F. Receiver (FIG. 5)

Before proceeding with a detailed description of the R.F. receiver 18 itis noted that while that receiver actually has a total of ten identicalchannels (a separate channel for each of the signals from each of thefour antenna quadrants 14A, 14B, 14C, 14D (FIG. 3A) and two auxiliarychannels which are time shared between the sidelobe cancelling subarrays(not shown) and the antenna segments 14AS, 14BS, only a pair ofchannels, corresponding to those passing the signals from antennaquadrant 14A, will be described in detail. It should be appreciated thatthe remaining channels are identical to those to be described.

Referring now to FIG. 5, the vertically and horizontally polarized inputsignals from circulators 251V, 251H (FIG. 4A), respectively, are appliedto a pair of bandpass filters 283V, 238H. Such filters, here waveguidedevices, are provided to reject undesired, out-of-band signals of anynature. Following the bandpass filters 283V, 283H are a pair ofwaveguide limiters 285V, 285H which are of conventional design and may,for example, comprise self-activated TR tube limiters. The latter areprovided to limit the input power to the R.F. receiver 18, therebypreventing burnout of the latter. Following the limiters 285V, 285H are:(a) a pair of variable PIN diode attenuators 287V, 287H; (b) a pair ofisolators 289V, 289H; (c) a pair of field effect transistor (FET)amplifiers 291V, 291H; (d) a second pair of isolators 293V, 293H; and(e) a second pair of PIN diode variable attenuators 295V, 295H, all ofwhich are of conventional design and are here fabricated in stripline.Each of the variable attenuators 287V, 287H, 295V, 295H (all of whichare controlled by control signals provided by the radar synchronizer 34(FIG. 1), has a 40 dB dynamic range. The inclusion of the variableattenuators 287V, 287H prior to, and the variable attenuators 295V, 295Hafter the FET amplifiers 291V, 291H allows spatial filtering, i.e. threephase center monopulse filtering to be described hereinafter, to beemployed as a counter-countermeasure against an escort jammer and adelta jammer. The attenuators in such a tactical situation preventsaturation so that the requisite linearity of response is maintained toallow spatial filtering to be carried out in a proper manner. It isnoted here that the bandwidth of the R.F. receiver being described isone-half the first I.F. frequency. This relationship then inhibits theappearance of a false I.F. signal from a delta jammer. A bandpass filter303 is provided in front of the mixer 305 to protect the latter fromspurious cross products which may be generated when the FET amplifiers291V, 291H are forced into saturation.

The output signals from the variable attenuators 295V, 295H are passed,via phase shifters 297V, 297H, respectively, to a power combiner 299wherein they are combined to provide a single output signal. Thevariable attenuators 295V, 295H and phase shifters 297V, 297H are, aswill be explained hereinbelow, includes to provide the radar system 10(FIG. 1) with radome compensation capability. The output signals fromthe power combiner 299 are passed, via a directional couple 301 and thebandpass filter 303, to the mixer 305 wherein they are heterodyned withthe first L.O. signal from the exciter 32 (FIG. 2), ultimately toproduce an S-band first I.F. signal. A preamplifier 307 is provided toamplify the S-band signals from the mixer 305 as such signals are passedto the I.F. receiver 20 (FIG. 1). The directional coupler 301 isincluded to allow injection of the REPLICA of the transmitted signal(obtained from the power distribution network 281 (FIG. 4B)) into thereceiver channel (not numbered). The REPLICA is processed and stored inthe signal processor 26 (FIG. 1) prior to the reception of any echosignals.

It is here noted that the REPLICAs of the transmitted pulses are passedthrough A/D converters after correlation in the I.F. receiver 20(FIG. 1) with the second L.O. signal from the exciter 32 (FIG. 1). Theresultant digital signals therefore represent the phase of the L.O. atthe time of transmission and the phase errors due to differentialtransmission line perturbations between the antenna 14 (FIG. 1) and thesecond correlation mixer. After T·B samples (where B is the A/D samplingrate an T is the pulse duration) are taken of the REPLICAS, theresulting number is stored within the signal processor 26 (FIG. 1) andis used to represent a phase reference for all received signalsresulting from that particular pulse transmission. Therefore, the signalprocessor 26 (FIG. 1) may cause return signals to be corrected for theeffects of phase modulation and phase distortion as mentionedhereinbefore.

As also mentioned hereinabove, phase shifters 297V, 297H are includes toprovide the radar system 10 (FIG. 1) with a radome compensationcapability. Before proceeding with a detailed description of thiscapability the need for a radome compensation technique will first beexplained. Thus, the radar system 10 (FIG. 1) employs a so-called "R.F.Cueing Mode" for the detection, location and subsequent attack ofground-based threat radars. In this mode the radar system 10 (FIG. 1)must passively detect a source of R.F. energy (a threat radar) andlocate the source both in angle and range so that a high resolutionsynthetic aperture radar map may be made of the surrounding area. Aftermapping of such surrounding area, the radar system 10 (FIG. 1) must thentrack the threat radar so that an attack may be mounted and carried outeven when the threat radar ceases to transmit or to scan.

In order to initiate tracking, accurate directional information of thethreat radar must be obtained (for example, by using the antenna 14(FIG. 1) as passive sensor) and the Doppler shift of return signals fromthe cell in which such radar is located derived. The requisitedirectional information (in elevation and azimuth relative to theaircraft carrying the radar system 10 (FIG. 1) determines the initialposition to which the antenna 14 (FIG. 1) is to be slued to commence mapmaking of the proper area of the ground. The Doppler shift of thesignals from the cell at such initial position is, of course, a functionof aircraft velocity and the directional information. With the antenna14 (FIG. 1) slued to the initial position, the radar system 10 (FIG. 1)is actuated to transmit. Range strobes are then used to sweep in rangeuntil the range to the center of the elevation monopulse null ismeasured. The radar system 10 (FIG. 1) then commences mapping the area(here to a 20 square foot resolution), measuring the azimuth andelevation angular location of each cell in the vicinity of the mapcenter as the mapping is effected. For each cell the azimuth andelevation angles, the range and Doppler shift are recorded.Consequently, a file of data is built up, defining ground reflectors inthe cells about the line-of-sight (LOS) to the threat radar. Withknowledge that mapping was initiated when the antenna 14 (FIG. 1)pointed directly at the threat radar, the cell at the intersection ofthe azimuth elevation nulls is first assumed to be the cell in which thethreat radar is located. It will be appreciated, however, thatsituations may exist when return signals from adjacent cells may be sosimilar to the returns from the first assumed cell that the threat radarcould very well be located in one of such cells. In such a situation,the cell with the return signals having the largest signal-to-noiseratio is selected. In any event, the range and Doppler coordinates ofthe selected cell are used to initiate a Doppler tracking loop (notshown) and the azimuth and elevation tracking loops (also not shown).

It should now be apparent that the accuracy of the foregoing threatradar tracking technique is dependent upon knowing the precise angularlocation of a ground target. To put it another way, the azimuth andelevation of the threat radar must be determined to within 0.50milliradians (mrad) to provide a cross-range accuracy of 15 meters at a30 Km range. Therefore, the effects of radome induced errors must beconsidered because radome boresight errors resulting fromcross-polarization and refraction effects can introduce up to ±20 and ±8mrad of error, respectively.

Digressing here now for a moment, it will be appreciated by those ofskill in the art that, in general, in a phase monopulse radar system theΣ channel signal is formed by vectorially adding the received signals inthe four quadrants while the Δ channel signals are formed by vectoriallysubtracting the received signals in opposite halves of the aperture ofthe antenna. After limiting, or otherwise equalizing, the amplitude ofthe received signals prior to forming the Σ and Δ signals, the quotientof Δ/Σ is a measure of boresight error which is independent of signalamplitude. However, any phase error introduced into the received signalsprior to forming the Σ and Δ signals will cause a boresight error.

In order to understand the effect of radome cross-polarization onboresight error, a short discourse on the nature of polarized signals isin order. In general, a polarized microwave ray passing through a radomeat an incident angle (i.e., and angle not perpendicular to the surfaceof the radome where the ray strikes), can be resolved into apolarization component parallel to the plane of incidence (the planebetween the incident ray and the normal to the radome surface) and apolarization component perpendicular to the plane of incidence. Theradome passes these two components with different degrees of attenuationand different microwave phase delays with the result that the rayemerging from the inside of the radome on the antenna has had itspolarization altered. If the receiving antenna was originallyco-linearly polarized with the impinging ray and the polarization ofthat ray was shifted by the radome, a component of the cross-polarizedenergy will be accepted by the antenna. Since the cross-polarized energyhas a different microwave phase than the co-polarized energy, theresulting vector will be phase-shifted and will create a boresight errorwhen processed.

For purposes of explanation, visualize two microwave rays of the samepolarization impinging on a radome at equal and opposite angles ofcurvature (as they would when entering the two halves of a monopulseantenna). It is also assumed that the polarization of the two rays isneither parallel to, or orthogonal to, the plane of incidence where theyenter the radome. For radome transmission purposes, the polarization ofeach of the rays may be resolved into a component parallel to the planeof incidence and a component orthogonal to the plane of incidence. Thetransfer function through the radome for each resolved component thenresults in differing signal vectors which, when combined, define themicrowave signal after passage through the radome. The apparent originof such a signal ordinarily differs from the actual origin of themicrowave signal being processed, i.e. there ordinarily is a boresighterror.

The two prime methods of compensating for radome induced boresighterrors are either: (a) to apply a correction factor to the boresighterror, or (b) to compensate each monopulse quadrant signal with aninverse radome transfer function. To correct the boresight errorrequires a knowledge of the signal polarization which cannot bedetermined until the signal is processed. The contemplated inversetransfer function method of compensation was chosen because it can beapplied to signals of any polarization without a priori knowledge ofthat polarization.

The inverse transform method of radome calibration requires a monopulseantenna in which each quadrant is sensitive to two orthogonalpolarizations. A microwave attenuation and phase shifting network isconnected to each of the orthogonal ports of each quadrant. The transferfunctions of these networks are digitally set to the mathematicalinverse of the radome/antenna transfer function for the associatedpolarization and quadrant. Since this transfer function changes with,inter alia, the incident angle of the signal upon the radome, amultiplicity of commands must be stored in order to cover all possibleincident angles so that the residual boresight error after compensationnever exceeds 0.5 milliradians.

Referring briefly now to FIG. 5A, a flow path in mathematical form for aparticular quadrant is illustrated to show how compensation is to beeffected in the contemplated way. The radome/antenna transfer functioncan be represented as a two-by-two matrix which multiplies theorthogonally polarized signals passing through the radome to reach theantenna quadrant. The method of compensation illustrated in FIG. 5A thenperforms an inverse matrix to the one representing the effect of theradome to produce two compensated signals which are identical to theones which entered the radome for that particular antenna quadrant. Itshould be noted here that in the radar system 10 (FIG. 1) completepolarization information is not maintained after the compensationprocess is completed since this would require ten receiver channelsinstead of five. A selection is made of a particular or desiredpolarization (say for purposes of rain rejection) by adding together thehorizontal and vertical components as illustrated in FIG. 5A.

Referring briefly now to both FIGS. 5 and 5A, the contemplated radomecalibration technique will be described. It will be understood, ofcourse, that the contemplated technique may be carried out on anyconvenient type of radar test range where the desired polarizations andfrequencies of microwave energy may be directed toward an assembly to becalibrated and the orientation (relative to the source of the microwaveenergy) of the radome in such assembly may be adjusted as required.Before proceeding, however, it should be noted that the calibrationprocedure will be described with reference to the receiver channel (notnumbered) dedicated to antenna quadrant 14A and that the calibrationprocedure would be repeated for each one of the remaining channels (notshown).

As mentioned hereinabove, the degree of randome cross-coupling isdependent, inter alia, on the sense of the incident polarization, theoperating frequency and the antenna position. Consequently, a verticallypolarized signal is first transmitted toward the antenna 14 (FIG. 1)with the radome being calibrated in place so that the response of theassociated receiver may be measured and recorded. Next, keeping theoperating frequency and antenna position constant, a horizontallypolarized signal is transmitted and the response of the associatedreceiver may again be measured and recorded. The process issystematically repeated, changing the orientation of the radome beingcalibrated and the frequency of the microwave energy so that theresponse of the associated receiver to any microwave energy underdifferent operational conditions may be measured and recorded.

It will be instructive at this point to recall that a polarizedmicrowave ray passing through a radome at an incident angle (i.e., anangle not orthogonal to the surface where the ray strikes) may beresolved into a polarization component parallel to the plane ofincidence (the plane defined by the incident ray and the normal to theradome surface) and a polarization component perpendicular to the planeof incidence. The two components are passed through the radome withdifferent degrees of attenuation and different microwave phase delays,with the result that the polarization and amplitude of a ray emergingfrom the inside of the radome differs from an incident ray.

Referring now to FIG. 5A, a radome compensation flow path inmathematical form for a particular antenna quadrant is illustrated.Thus, the effect of the radome may be mathematically represented as atwo-by-two matrix which multiplies orthogonally polarized signals whichpass through the radome to reach the antenna quadrant. The compensationcircuitry (not numbered) performs an inverse matrix to the onerepresenting the effect of the radome and, therefore, two compensatedsignals are produced which correspond with the incident orthogonallypolarized signals.

With the foregoing in mind, it is here contemplated that each radome andantenna assembly be calibrated on a range where the orientation of suchassembly to a linearly polarized test signal may be measured and theoperating frequency of such a signal may be controlled to deriveappropriate compensation signals for any orientation of the radome andantenna assembly with respect to any incident signal at any frequencywithin the operating bandwidth of such assembly. Thus a radome andantenna assembly to be calibrated is positioned in any convenient manneron a range to receive a linearly polarized test signal, E, of the form(simplified)

    E=E.sub.i e.sup.jωt (cos θ+sin θ)

where E_(i) is the amplitude of the test signal and θ is thepolarization angle. It will be recognized that such a signal is theequivalent of two orthogonally polarized signals, one (the cosinecomponent) being horizontally polarized and the other (the sinecomponent) being vertically polarized. Each such component, upon passingthrough the radome and antenna assembly under test is, in turn,subjected to differing degrees of attenuation and phase delays. Thus,after the test signal passes through the radome and antenna assembly theoriginal horizontally polarized component may be expressed as

    A.sub.H =E.sub.1 (G.sub.XX cos θe.sup.j(ωt+D.sbsp.1.sup.) +G.sub.XY sin θe.sup.j(ωt+D.sbsp.2.sup.))

where G_(XX) and G_(XY) are attenuation factors and D₁ and D₂ are phaseshifts.

Similarly, for the original vertically polarized component,

    A.sub.V =E.sub.1 (G.sub.YY sin θe.sup.j(ωt+D.sbsp.4.sup.) +G.sub.YX cos θe.sub.j(ωt+D.sbsp.3.sup.))

The attenuation factors and phase shifts describe the transfer functionof the radome and antenna assembly under the given conditions.

The calibration procedure then is directed to the calculation of theinverse radome and antenna transfer function for the given conditions.

The determinant G [ ] associated with the radome compensation network(not numbered) is of the form:

    G[ ]=G.sub.XX G.sub.YY e.sup.j(D.sbsp.1.sup.+D.sbsp.4.sup.) -G.sub.XY G.sub.YX e.sup.j(D.sbsp.2.sup.+D.sbsp.3.sup.).

From the foregoing it should now be apparent that if G_(H) =cos θ andG_(V) =sin θ, the resultant signal from a radome compensation andpolarization selection network should be A_(i) e^(j)(ωt).

The contemplated calibration technique will now be described in terms ofthe mathematical model of FIG. 5A. Thus, as mentioned above, thecalibration procedure for a given antenna look angle, antenna rollangle, and operating frequency involves first illuminating the radomewith a horizontally polarized signal and measuring the response, andnext, keeping everything else constant, illuminating the radome with avertically polarized signal and measuring the response to this input.The detailed procedure is as follows: With a horizontally polarizedinput signal variable attenuators 287V and 295V (FIG. 5) are set toprovide maximum (80 dB) attenuation (in effect turning OFF thevertically polarized channel (not numbered)) and the response as seen bythe signal processor 26 (FIG. 1) is recorded. This response correspondsto the G_(XX) e^(j)(D.sbsp.1.sup.) term in mathematical model of FIG.5A. Next, still with the horizontally polarized input signal, variableattenuators 287H and 295H (FIG. 5) are set to provide maximumattenuation (thereby turning OFF the horizontally polarized channel)while variable attenuators 287V, 295V are set to provide minimumattenuation and the response as seen by the signal processor 26 (FIG. 1)is recorded. This response corresponds to the G_(YX)e^(j)(D.sbsp.3.sup.) term. The foregoing procedure is repeated with avertically polarized input signal to yield the G_(YY)e^(j)(D.sbsp.4.sup.) and G_(XY) e^(j)(D.sbsp.2.sup.) terms. On the basisof these measurements, the determinant G[ ] and the terms ##EQU1## arecomputed and stored in memory.

Digressing briefly here now for a moment, it should be noted that thevariables considered here to be important to radome compensation areantenna look angle, antenna roll and operating frequency. Given thevariables selected, it may be estimated that a solid angle of 0.03steradians (10°×10°) would encompass a volume within which the boresighterror would be permissible and that a total of eight roll positionswould cover the ±75° of roll required by a 4 g accelerationspecification. Twenty frequencies were selected to cover the operatingbandwidth of the system. Based on the foregoing, a total ofapproximately 4×10⁶ bits of memory are required for the requisite datafor radome calibration. Further, it should also be noted here that thetransfer function through the radome (not shown), the antenna 14(FIG. 1) and the R.F. receiver 18 (FIG. 1) may be represented solely bya network comprising a pair of attenuators and a pair of phase shifters.Thus, the radome compensation is provided through the use of a pair ofvariable attenuators and a pair of phase shifters in each channel of theR.F. receiver 18.

Proceeding now with the description of the calibration technique, onceall the constants are calculated and stored, the procedure is to selecta desired polarization (for a given operating frequency, antenna lookangle and antenna roll angle), calculate the requisite transferfunction, and convert the calculated transfer function into equivalentattenuator and phase shifter values.

It is noted here that the tolerances of the attenuators 295V, 295H andthe phase shifters 297V, 297H are relaxed because the pilot pulse isused to set them to the correct value.

It should be noted here that the contemplated radome calibrationtechnique also provides an advantage in terms of two way antennasidelobe control with a nonreciprocal antenna. That is to say, ontraversing the radome a transmitted beam will have impressed on it asidelobe pattern dependent on the attenuation and phase delay throughthe radome. Without compensation, the corresponding received beam would,upon traversing the radome, have an entirely different sidelobe patternimpressed on it, depending upon the motion of the aircraft between thepulse transmission and reception times. The contemplated radomecompensation technique would, however, remove the effect of the radomeon the received beam with a concomitant improvement in operation.

I. F. Receiver (FIG. 6)

Referring now to FIG. 6, a single channel of the I.F. receiver 20,corresponding to that associated with antenna quadrant 14A, will bedescribed in detail. It should be appreciated that the remainingchannels (not shown) are identical to that to be described. The S-bandI.F. signals from the R.F. receiver 18 (FIG. 5) are passed, via anisolator 311, through a bandpass filter 313 to a correlation mixer 315,which also is fed by the second L.O. signal from the exciter 32 (FIG.2A) to down-convert the S-band I.F. signals to a second I.F. frequencyof 100 MHz. It should be noted here that the correlation mixer 315 alsoperforms the correlation or "stretch" processing for the chirpedwaveforms. The second I.F. signal from the correlation mixer 315 isamplified by an amplifier 317 and split into two channels (notnumbered), the upper channel being used in the short range tracking modeand the lower channel being used in the mapping and long range trackingmodes. It should be noted here that two channels are provided because ofthe different dynamic range requirements which are imposed on the twotracking modes.

The lower channel (sometimes hereinafter referred to as the mapping andlong range tracking channel) comprises an amplifier 319, a limiter 321L,an amplifier 323L and a bandpass filter 325L, all of which are ofconventional design. The upper channel (sometimes hereinafter referredto as the short range tracking channel) includes a limiter 321U, andamplifier 323U and a bandpass filter 325U, all of which are also ofconventional design. The bandpass filters 325U, 325L are provided toremove any harmonics or spurious signals generated within the limiters321U, 321L and the amplifiers 323U, 323L.

The filtered output signals from the bandpass filters 325U, 325L aresplit and applied to mixers 327UI, 327UQ, 327LI, 327LQ wherein they aredownconverted to baseband quadrature video signals by being heterodynedwith COHO L.O. signals obtained from the exciter 32 (FIG. 2A). It shouldbe noted that the requisite quadrature relationships are obtained bypassing the COHO reference signals to mixers 327UQ, 327LQ through 90°phase shifters 329U, 329L, respectively. The output signals from mixers327UI, 327UQ, 327LI, 327LQ are filtered by low pass filters 331UI,331UQ, 331LI, 331LQ to remove undersired sidebands and are amplifiers byamplifiers 333UI, 333UQ, 333LI, 333LQ prior to being passed to switches335I, 335Q. The latter, which are controlled by control signals providedby the radar synchronizer 34 (FIG. 1), are effective to gate either thelong or short range tracking waveforms through a video receiver (notnumbered). The long range tracking waveforms are obtained from themapping and long range tracking channel wherein they are subjected to anadditional stage of amplification by virtue of the amplifier 319.

The selected waveforms from the switches 335I, 335Q are sampled andgain-scaled in sample/hold circuits 337I, 337Q, of conventional design,by mode control signals from the radar synchronizer 34 (FIG. 1). Thesampled output signals from the sample/hold circuits 337I, 337Q aredigitized by ten bit A/Ds 339I, 339Q and are passed, via buffers 341I,341Q, to the signal processor 26 (FIG. 1).

The mapping signals in the long range track and mapping channel (notnumbered) are amplified in buffer amplifiers 343I, 343Q prior to beingapplied to eight bit A/Ds 345I, 345Q. The latter have a throughput rateof 50 MHz which is faster than the throughput rate of the signalprocessor 26 (FIG. 1). In consequence, then, "first in, first out"memories (FIFO 349I, 349Q) are provided to store a series of real timesamples from the A/Ds 345I, 345Q and provide output signals to thesignal processor 26 (FIG. 1) at a slower data rate. To reduce thestorage requirements of the FIFO memories 349I, 349Q, digital filters347I, 347Q are provided between the A/Ds 345I, 345Q and the FIFOmemories 349I, 349Q. The digital filters 347I, 347Q perform anintegration (or summation) of several contiguous samples which appear atthe output of the A/Ds 345I, 345Q. The number of outputs to be summedbefore being dumped into the FIFO memories 349I, 349Q is determined by acontrol signal provided by the radar synchronizer 34 (FIG. 1).

Completing the I.F. receiver 20 are a pair of phase shifters 351U, 351Land a phase shifter control unit 353, which is shown to receive inputsignals from the radar synchronizer 34 (FIG. 1) as well as from the fivevibration sensitive piezoelectric accelerometers (not shown) on the rearface of the antenna 14 (FIG. 1). The phase shifter control unit 353,which here is simply a double integrating filter, is effective todevelop control signals for the phase shifters 351U, 351L, dependingupon the vibration experienced at each antenna quadrant phase center attime of pulse transmission as well as the frequency of the transmittedpulse. The piezoelectric accelerometers (not shown) are provided tomeasure the differential vibratory motion between the antenna phasecenters, corresponding to the centers of antenna quadrants 14A . . . 14D(FIG. 3A), which, unless compensated for, would result in phase trackingerrors between the latter. Thus, the phase shifters 351U, 351L areprovided to compensate for any differential vibratory motion between theantenna quadrants 14A . . . 14D (FIG. 3A) by rotating the COHO signalbefore it is applied to the quadrature downconversion mixers 327UI,327UQ, 327LI, 327LQ.

It is anticipated that the vibratory motion of the antenna willintroduce approximately ±6 degrees of phase error in the receivedsignals and, therefore, the design of the phase shifters 351U, 351Lbecomes less critical. Thus the latter may, for example, be fabricatedby terminating the output ports of a quadrature hybrid with varactordiodes. As the impedance of the diodes is changed in response to anapplied voltage the phase of the signal reflected from the output portswill be correspondingly altered.

Radome Compensation Electronics

As hereinbefore described, the compensation for radome error requiresthe adjustment of two compensation elements (variable attenuators 295V,295H and phase shifters 297V, 297H being shown in FIG. 5) in eachchannel of the R.F. receiver 18 (FIG. 1). This means that a total ofthirty-two compensation elements must be digitally controlled. The bitlength is a maximum of 10 bits for any one element. Further, becauseeach control word must be updated whenever frequency or antenna pointingdirection is changed and because the calculation of each control word isperformed in the radar synchronizer 34 (FIG. 1), an interface betweeneach compensation element and the radar synchronizer 34 (FIG. 1) isneeded. The requisite interface is provided by an interface in the radarsynchronizer 34 (FIG. 1), which interface is similar to the one shown inFIG. 9B. Because the construction of the interfaces are so similar, onlyone will be illustrated.

To allow asynchronous operation, two memories are required, one storingcontrol signals from the radar synchronizer 34 and the other providingcontrol signals to a set of thirty-two buffer registers. The memoriesperform the so-called "corner turning" function to convert each controlsignal to a serial form for entry in the proper buffer register. Eachone of such registers converts the applied control signal back to aparallel form and holds that signal for the associated compensationelement.

The requisite control signals are partially calculated in the radarsynchronizer 34 (FIG. 1). Common inputs for the calculation of all ofthe control signals include frequency and antenna pointing directionwhich together determine the address of a block of radome compensationdata stored on a dedicated disk. In addition, signals indicative oftemperature of the antenna 14 (FIG. 1), radar mode and weatherconditions are supplied to digital data processor 30 (FIG. 1) forcompleting the calculation of the radome compensation values.Combinations of the addressed block in the dedicated disk and theoutputs of the digital data processor 30 (FIG. 1) then make up thecontrol signals.

A complete data set for the radome compensation process consists ofthirty-two control signals for each of the four frequencies transmittedafter each PRI. Assuming a 10 bit digital word for each control signal,there are 1280 bits in a complete data set. Although updating here ofthe control signals in only one quarter (8) of the buffer registers needbe done when frequency is changed, it is preferred here to update all ofsuch buffers when frequency is changed. The use of two memories makessuch complete updating possible without interfering with calculation ofnew control signals and eliminates the necessity for differing updatingmodes when, for example, updating is required when the pointingdirection of the antenna is changed with respect to the randome.

Because compensation must be done in real time, double buffering at eachcompensation element is dictated to allow each one of the thirty-twocontrol signals to be changed four times in each PRI (with the maximumallowable time for change being one microsecond). Double buffering ateach compensation element lengthens the maximum allowable time forchange to twenty-five microseconds without affecting performance. Justafter a pulse is transmitted and before the R.F. receiver 18 (FIG. 1) isenabled, each newly assembled set of control signals then is clockedinto a corresponding double set of double buffer registers 298 (FIG. 5).

The provision of two memories and the sets of double buffer registers298 (FIG. 5) relieves the digital data processor 30 (FIG. 1) of veryhigh overhead real-time refresh chores.

Within the radar synchronizer 34 (FIG. 1) the memory used for storingthe control signals is arranged so that one bucket of such memory isloaded while the corresponding bucket of the other is clocked out. Therequisite bucket switching (which is conventional) is performed whenneeded under control of the digital data processor 30 (FIG. 1), at mostevery 10 ms (during a worst case roll).

It is noted here that the bandwidths and center frequencies of the I.F.receiver 20 are designed to be immune to delta jamming by using the sametechnique as was described in connection with the description of theR.F. receiver 18. That is to say, the bandwidth of each section of theI.F. receiver 20 is designed to be one-half the frequency of eachsucceeding downconverter so no false I.F. signal may be engendered fromthe difference signal of a delta jammer.

Motion Compensation (FIGS. 7A and 7B)

Before proceeding with a detailed description of the radar synchronizer34 (FIG. 1), the signal processor 26 (FIG. 1) and the digital dataprocessor 30 (FIG. 1), it will be instructive at this point to discussthe motion compensation requirements of the radar system 10 (FIG. 1).

As is known, any synthetic aperture radar used to map a terrain withhigh resolution utilizes coherent processing of signals from eachindividual ground reflector in the terrain being mapped for a dwell (orcoherent integration time) of sufficient length to resolve such signalsin a set of range gates and Doppler filters. Such processing makesdirect use of the motion of an airborne radar during the dwell period toprovide different Doppler frequencies for ground reflectors at differentangles from the velocity vector of an aircraft. However, significantmovement of the range gates and Doppler filters during the dwell inducequadratic and cubic phase components on the terrain and causedegradation of the resolution of the map finally produced. Compensationfor such movements includes "slipping" range gates and the Dopplerfilter bank with sufficient accuracy in a known manner to reduce theunwanted motion during each dwell to acceptable bounds.

A further source of error in radar mapping is phase modulation, inducedby random or vibrational motions of the phase center of the radarantenna, during each dwell. With such motions, if the resulting phasemodulation is sufficiently large, modulation sidebands of signalsassociated with each particular ground reflector may overlap withsignals from nearby ground reflectors. Compensation for such motionstherefore consists of impressing a time-varying phase shift on eachsignal being processed, such phase shift being determined by a sensormonitoring the phase center motion of the radar antenna along boresightand being opposite in sense to the phase modulation introduced by theunwanted motion of the radar antenna.

The Doppler frequency of any ground reflector will change during eachdwell as a result of the slightly changing angle to each groundreflector caused by the motion of the radar antenna. If such change issufficiently large during each dwell so that the Doppler resolutionwidth is exceeded, then a spreading of the energy from ground reflectorsinto adjacent Doppler filters results. To prevent such spreading,compensation in the form of an appropriate time-varying Doppler shift ofeach received signal is required. In this case, however, inertialinstrumentation is used to sense the motion causing the unwanted changein Doppler frequency.

In addition to the foregoing, other perturbations of the antenna causephase modulation of the received signals. If these modulation terms arein a small bandwidth, i.e., small with respect to the Doppler resolutionbandwidth (which is approximately equal to the inverse of the time ofdwell), then the effect is apparently the same as a Doppler modulationcaused by a slowly moving ground reflector, with a maximum Doppler rategiven by:

    f(t)=2a(t)/λ                                        Eq. (1)

where a(t) is the acceleration of the antenna phase center in thedirection of the line-of-sight and λ is the wavelength of thetransmitted signals. When modulation containing significant energy atfrequencies higher than the coherent resolution bandwidth are caused, aphase error that varies during each dwell, a results. If such errorchanges sinusoidally during a dwell, a spurious pair of sidelobes in theDoppler filter output is created. The Doppler offset of such sidelobes(relative to the mainlobe filter response) is equal to the frequency ofthe perturbation causing the unwanted phase modulation. The amplitude ofeach such sidelobe (relative to the mainlobe filter response) isdirectly proportional to the peak excursion of the phase center of theradar antenna along boresight.

Conventional motion compensation techniques cancel the effects of radialacceleration from the radar return signals. However, under dynamicflight conditions as here contemplated, cancellation of such effects isnot of prime concern. Rather, the most important object is to attainwidth and depth of focus of any map when violent maneuvers are requiredin a tactical situation without exceeding the computational timeconstraints. It can be shown that the horizontal component of aircraftacceleration perpendicular to the antenna boresight direction (usuallydenoted simply as the cross-acceleration) has a devastating effect onthe width of focus when any conventional motion compensation techniqueis followed.

In order to maintain the width, depth and "height" of focus of therequisite synthetic aperture maps under severe aircraft acceleration,the pulse repetition interval (PRI) of the herein contemplated radarsystem 10 (FIG. 1) is varied to hold constant the angular incrementswept out by the horizontal component of the line-of-sight between theantenna and the map reference point. The width of focus then is afunction of the horizontal component of radial acceleration as opposedto the cross-acceleration. The depth of focus, on the other hand, is afunction of both vertical and horizontal radial acceleration components.The depth of focus is, however, not as strongly dependent on aircraftacceleration as is the width of focus, and the depth of focus isrelatively independent of PRI variation.

"Height" of focus refers to the height (above or below) that a givenground reflector may be (with respect to that of a map reference point)without undue defocusing. The height of focus is limited by thehorizontal (radial) and vertical components of aircraft acceleration andcentripetal acceleration effects, with the vertical component of vehicleacceleration being the most important. Since the height of focus isproportional to the square of the azimuth resolution, it imposes asevere restriction when a map with very high resolution is to be made.The height of focus cannot be increased in any direct manner by moreprocessing capability since it is jointly dependent upon vehicleacceleration and terrain variation.

Referring now to FIGS. 7A to 7C, the effects of aircraft accelerationson the problem of focusing a synthetic aperture map will be explained.Thus, an aircraft 302 (FIG. 10) employing a synthetic aperture radar(not shown) has its radar antenna (also not shown) located at point a,while a synthetic aperture map reference point is located at point o. Aground scatterer is located at a point i with cartesian coordinatesX_(i), Y_(i), Z_(i) with respect to the map reference point o. Areference coordinate frame (the X_(o), Y_(o) and Z_(o) coordinate frame)is a cartesian coordinate frame centered at the map reference point o,with the Z axis being vertical and Y axis being along the horizontalcomponent of the position vector (not numbered) from point a to point oat time t_(o). The X axis completes the orthogonal triad, here in therighthanded sense. The X₁, Y₁, Z₁ coordinate frame is another localvertical frame centered at point a, with the Z_(I) axis being verticaland the Y_(I) axis disposed along the horizontal projection of thevelocity vector of the aircraft 302 (FIG. 10). A_(o) is the anglebetween the line-of-sight vector to the map reference point o and thevelocity vector. The angles B_(o) and C_(o) are, respectively, thesquint and elevation angles.

It will be noted that, as the aircraft 302 (FIG. 10) moves, theboresight line of the antenna remains fixed on point o, which means thatthe X_(o),Y_(o) coordinates of the point o rotate about the Z_(o) axisand that (for small changes of position of the aircraft 302 (FIG. 10))the X_(i),Y_(i) coordinates of the point i similarly rotate about theZ_(o) axis.

It will be apparent that no cross-acceleration, i.e. acceleration alongthe X axis, resulting from movement of the aircraft 302 (FIG. 10) isexperienced at the point o. It follows then that any compensationtechnique for nonlinear motion between the aircraft 302 (FIG. 10) andthe point o need only be directed to compensation for radialacceleration, i.e. acceleration along the Y axis. On the other hand,because of the displacement of the point i from the point o, motion ofthe aircraft 302 (FIG. 10) causes (in addition to a radial movementalong the Y axis), cross-acceleration along the X axis and a concomitantnonlinear change in phase of the returned signals from the point i. Themagnitude and sense of the cross-acceleration (and also of the radialacceleration) at the point i is a function of its direction anddistance, measured in the X-Y plane, from the point o. As mentionedhereinbefore, compensation for the displacement of the point i along theZ axis may not be directly calculated; however, if vertical accelerationof the aircraft 302 (FIG. 10) is reduced to a minimum during a dwell,the compensation for cross-acceleration and radial acceleration willalso provide a modicum of compensation for "Z" axis displacement. Inthis connection it will be recognized that an inherent ambiguity existsbetween return signals from a point above the plane of the selectedrange-Doppler matrix, i.e. the X-Y plane, and return signals from apoint below such plane. A moment's thought will make it clear that,although the selected range-Doppler matrix is defined in the X-Y planeby appropriate pairs of isorange and isodop lines, such pairs of linesare also lines which lie on curved surfaces. The isorange lines lie onspherical sectors of "range" spheres which are centered at point a andthe isodop lines lie on the surfaces of halves of "Doppler" cones havingvertices along the velocity vector and intersecting the terrain beingmapped to define the isodops as hyperbolas. Return signals from anypoint within the space between selected spherical sectors and the"Doppler" cones have the same "range-Doppler" signature and aretherefore undistinguishable from each other. Further, all such signalswould have the same "range-Doppler" signature as return signals from apoint on the Y coordinate. The result is that a three dimensional fieldmapped by radar appears as a two dimensional image in which it appearsthat there is no difference in elevation acrpss such field. Fortunately,however, there is little chance for confusion (when a terrain is beingmapped) as to whether a given target is "higher than" or "lower than"the point o. On any terrain being mapped, all targets of interest areground targets approximately on the same level as the center of the map,here point o. Further, because of the long experience people have hadwith conventional photography, the interpretation of images in twodimensions of features which are actually three dimensional is easilyaccomplished.

It will now be appreciated that, if the differences between the radialacceleration at the points o and i and the cross-acceleration at thepoint i are determinable, then compensation for motion may be effectedfor points on the map other than point o. As a result, then, a radar mapof higher quality may be made. It will also be appreciated that thedegree of improvement, in a practical case, is the result of trade-offbetween the size and quality of the desired map and the necessity ofproviding a computer to calculate and to apply the requisitecompensation factors in real time. Obviously, it would be ideal tocalculate and apply a proper compensation factor to each range-Dopplercell in order to increase the resolution of each cell making up a radarmap. It is equal obvious, however, that the amount of data to beprocessed to obtain the large number of correction factors requiredwould make such an approach impractical, considering the present stateof the art of airborne computers. It is, however, here contemplated,using presently known computers, to calculate compensation factors toimprove resolution by at least an order of magnitude over the resolutionusually attained, making it possible to generate a radar map on whichsmall targets, such as tanks, may be distinguished.

With the foregoing in mind it is here contemplated to derivecompensation factors which improve the resolution of point i withoutallowing the nonlinear change in phase of signals from point o to exceeda given tolerable amount, say 45°. Because the phase of any receivedsignal is a function of the range of the reflector giving rise to suchsignal, the requisite compensation factors, i.e. phase shifts, may becalculated by first determining how the ranges to the points of interest(points o and i) change during a dwell.

Referring now to FIG. 7A, the geometry of an exemplary situation whereinan aircraft moves from point a to point b during a dwell (time interval(t₁ -t_(o))) while the boresight of an antenna (not shown) is kept onpoint o to measure the depression angle (which is the same as theelevation angle C when the aircraft is in level flight), the angle A andthe angle B and the range to point o. The aircraft here, in its inertialcoordinate system (X_(I), Y_(I), Z_(I)), has a velocity V, and issubject to acceleration which may be resolved into accelerationsA_(X)(I), A_(Y)(I), A_(Z)(I) along the inertial coordinates X_(I),Y_(I), Z_(I). In addition, it should be noted that relative movementsbetween the antenna 14 (FIG. 3A) and the aircraft 302 (FIG. 10) add tothe accelerations. It will be apparent from inspection of FIG. 7A thatthe range R_(i) to any selected points, here points o and i, on the X-Yplane may be determined and that the amount and direction of rotation ofthe X-Y axes may also be determined. To put it another way, using pointo as the reference point during a dwell, the position and orientation ofthe isorange lines and isodops in the "range-Doppler" matrix relative tothe inertial coordinates X_(I), Y_(I) may be determined. As a result,the signals returned from different points on the terrain being mappedmay be appropriately modified to allow the finally generated radar mapto be presented in inertial coordinates with greater resolution thanwould be possible if only point o were to be compensated.

It will now be appreciated that the rationale of the method of operatinga pulse radar disclosed by Nathan Slawsby in U.S. Pat. No. 4,084,158(which patent is assigned to the same assignee as this application) isapplicable here. That is to say, because the Doppler shift of thesignals returned from each point on the terrain being mapped changes ina nonlinear manner, i.e. exhibit a Doppler acceleration as well as aDoppler velocity, compensation for such nonlinear changes may beeffected here by changing the pulse repetition frequency of the radar sothat the received signals from each point on the terrain are properlyphased with respect to each other to minimize the effect of Doppleracceleration.

Although the method described by Slawsby in the cited patent iseffective in compensating for movement of the radar to improve theresolution of a radar map, it is not completely satisfactory by itselfin all situations. In particular, when a moving target (such as a tank)is to be detected in the presence of "clutter", the signal-to-clutterratio obviously should be optimized. Here "clutter" is made up of thefeatures on the mapped terrain which allow the radar map to be made,such features being stationary.

Optimization of the signal-to-clutter ratio may be effected here byfollowing an iterative procedure whereby: (a) the antenna 14 (FIG. 3A)is oriented so that, with the boresight line intersecting the point o,the phase centers A', B' are in the plane defined by the boresight lineto the point o and the velocity vector of the aircraft 302 (FIG. 10);(b) the pulse repetition interval between successive pulses is adjusted(as described hereinbefore) to reduce the effect of acceleration on thereceived signals from point o to a minimum; (c) the difference betweenthe signals received in the quadrants 14A, 14B, integrated during adwell, are compared to a predetermined level; and the process isrepeated (with different pulse repetition intervals) until only theintegrated returns from moving targets are above the predeterminedlevel. As indicated in FIG. 7B, the optimum pulse repetition intervaloccurs when, in the interval between successive pulses, the phase centerB' moves into coincidence with the line between the phase center A' andthe point o existing when a preceding pulse was transmitted.

It will be appreciated on inspection of FIG. 7B that the just-describedmovement of the phase center B' (to the point marked B") may beaccomplished only if the antenna is rotated about both the Z_(I) and theY_(I) axes as the antenna is moved. It will also be appreciated that,because the transmitted signals are coherently modulated and thedistance dR in FIG. 7B may be measured, the phase shift of receivedsignals due to the propagation delay suffered by any signal movingthrough the distance dR is calculable so that compensation of such phaseshift may be effected. To put it another way, the phase centers A' andB' may (after movement of the latter to point B" and compensation) beconsidered to be at the same range from any stationary object at or nearthe point o. It follows then that subtraction of one of the demodulatedreceived signals from the other will effect cancellation in the samemanner as a conventional monopulse receiver would be processing receivedsignals in two opposite parts of a monopulse antenna.

A moment's thought will now make it clear that compensation for thedistance dR to make it seem that the phase centers A', B" are coincident(as far as return signals from a stationary object adjacent the point oare concerned) may, or may not, have a similar effect on return signalsfrom a moving target. Thus, the distance dR may be defined in terms of anumber of wavelengths of the transmitted signal; this in turn means thatthe phase shift of any received signal moving through the distance dRmay be expressed as 2 πn/λ, where "n" equals the distance dR expressedin terms of the number of wavelengths of the transmitted signal. It willbe recognized that the expression just made is the same as theexpression for Doppler "blind" speeds (where "n" is any integer greaterthan unity). Therefore, if return signals from a moving target are beingprocessed, the range rate of such target (which, of course, determinesthe phase shift impressed on such return signals over a period of time)controls the degree of cancellation suffered when the signals from thephase centers A', B" are processed. That is to say, whenever a movingtarget has a Doppler "blind" speed, the return signals from such targetwill appear to be the same as received signals from a stationary object,with the result that processing cancels such return signal.

To reduce the number of Doppler "blind" speeds, it is necessary toreduce the distance, in wavelengths, between the phase centers, i.e.,reduce the size of "n". Here, because the phase centers A' and B" arefixed in position relative to one another, it is contemplated to processthe return signals at phase centers C', D' to produce an equivalentphase center E', midway between phase centers A', B'. Thus, the returnsignals at phase centers C', D' are added and the resultant sum signalis divided in half to produce the equivalent phase center E. When theequivalent phase center E is coincident with the original line from A'to the point o (again by controlling the pulse repetition intervals androtating the antenna) the intervals between Doppler "blind" speeds areincreased. It follows then that "low speed" targets may be detectedusing received signals from phase centers A' and B' and "high speed"targets tracked using received signals from phase centers A' (or B') andE'.

It will now be noted that, although FIG. 7B and the discussion inconnection with FIG. 7B have been specifically directed to how thesignal/clutter ratio adjacent the point o would be enhanced, thesignal/clutter ratio in other areas could be enhanced simply by havingthe phase center B" fall on the original line of sight to a point otherthan point o. It will also be noted that considering FIGS. 7A and 7Btogether, in a particular tactical situation with known dimensions ofthe antenna, there are a sufficient number of measured or known anglesand distances to allow a complete solution of the problem to be made.

Integrated Radar/Inertial System (FIG. 7C)

Referring now to FIG. 7C, a simplified block diagram of the radar system10 (FIG. 1) illustrating the interplay between the various subassembliesfor the purposes of motion compensation is illustrated. Thus, the linelabeled "PIEZO" out of the antenna 14 represents the output signals fromthe piezoelectric accelerometers (not shown) which are passed, here viathe synchronizer 34, to the I.F. receiver 20 (FIG. 1) wherein they areused, in a manner to be described in detail hereinbelow, to compensatefor any vibratory motion of the antenna phase centers (not shown). Theoutput signal from a radial accelerometer 171 (FIG. 8B) as well as theoutput signals from the cross accelerometers (not shown) and the rategyros (also not shown) within the head mounted IMU 24 are passed inanalog form to the boresight sensor assembly 36 wherein they areprocessed ultimately to form motion compensation signals for the fourquadrant phase centers (not shown). Thus, in order to calculate theantenna axis components of the gravity vector a processor, referred toas the strapdown processor 173, within the boresight sensor assembly 36computes a direction cosine transformation which defines the angularorientation of the antenna axes relative to the north, east and localvertical axes determined by the inertial platform 38. The angular motionof the radar antenna axes is sensed by the three orthogonal rate gyros(not shown) within the head mounted IMU and their output signals areused in the strapdown processor 173 for high frequency (200 Hz) updatingof the direction cosine matrix. Low frequency updating of the directioncosine is, as will be described in detail hereinbelow, provided via atransfer alignment Kalman filter (not shown but within the inertialplatform 38) which utilizes the coordinate frame instrumented by theinertial platform 38 as a reference for the strapdown direction cosinetransformation matrix.

The strapdown processor 173 rotates the three accelerometer inputsignals, via the direction cosine matrix, into locally level coordinatesand integrates the equations of motion to obtain velocity outputsignals. The latter, after conventional lever arm corrections in thesignal processor 26 to account for displacement vector changes betweenthe inertial platform 38 and the antenna 14, are passed to the inertialplatform 38. The latter compares, in a transfer alignment Kalman filter(not shown), the velocity estimates from the strapdown processor 173with its own velocity estimates. The velocity differences are the resultof errors in the direction cosine matrix within the strapdown processor173 as well as in accelerometer and gyro measurement errors. The Kalmanfilter (not shown) is designed to estimate these errors which aresubsequently passed back, as corrections, to the strapdown processor173.

The output signals from the radial accelerometer 171 (FIG. 8B) areaccumulated in high speed motion compensation buffer registers (notshown, but each having twenty-one bits) within the boresight sensorassembly 36. The buffer registers are bumped at a rate corresponding tothe PRF to provide signals for the synchronizer 34, to be digitallyintegrated along with feedback signals from the signal processor 26 toform incremented motion compensation (phase correction multiplies) forthe radar signal processing. The horizontal cross-accelerometer data isalso accumulated in the boresight sensor assembly 36 and sent to thesynchronizer 34 to be used for PRI control. The component of verticalacceleration due to antenna motion is also passed from the boresightsensor assembly 36 to the synchronizer 34 for use, in a manner to bedescribed in detail hereinbelow, for ultra high resolution A/D samplingrate variation.

The signal processor 26 performs the following high speed motioncompensation functions: Phase rotation of the PRF samples from each ofthe four antenna phase centers, phase rotation required for cluttercancellation, and differential phase corrections for map range sectors.The digital data processor 30, on the other hand, performs low frequencycentripetal motion compensation and supplies tracking error commands tothe boresight sensor assembly 36.

Gimbal Structure and Antenna IMU (FIG. 8A)

Referring now to FIGS. 8A to 8B, the antenna gimbal structure 200 isshown to comprise a four axis gimbal system. The yaw gimbal axis 201 andthe pitch gimbal axis 203 are conventional, being required to point theantenna boresight axis 205 in a given direction relative to the aircraftcenterline 207. In order to meet the requirement that the plane (notshown) containing the antenna azimuthal phase centers be maintainedparallel to the slant plane (the plane defined by the velocity vectorand the line-of-sight to the map reference point) a tilt (or elevation)gimbal axis 209 is required. It should also be noted that the slantplane is also the plane which cuts the isodop-isoran lines (not shown)at 90° and that, in consequence, the plane for the azimuth monopulseaxis, which is coincident with the plane (not shown) containing theantenna azimuthal phase centers, is also coincident to the slant plane.Finally, a roll bulkhead gimbal axis 211 is provided to prevent gimballock and to maintain the requisite tracking accuracy in the presence ofhigh angular accelerations.

It is noted than an inertial measuring unit 24 (sometimes hereinafterreferred to simply as the head mounted IMU 24) is provided on the gimbalstructure 200. The herein comtemplated system utilizes both the headmounted IMU 24 and the inertial platform 38 for the purposes of motioncompensation and navigation. If both functions were to be performed bythe head mounted IMU 24, a precision strap-down system with itsattendant performance degradations and computational difficulties wouldbe required. On the other hand, if the inertial platform 38 were to beused for both the motion compensation and navigation functions, itslocation within the aircraft 302 presents a problem. That is to say, fornavigation purposes it is desirable to locate the inertial platform 38at, or within the proximity of, the center of gravity of the aircraft302. This location here is about eighteen feet from the nose of theaircraft 302 where the antenna 14 is located. Since the instantaneousinertial acceleration at the antenna phase centers is not wellcorrelated with the inertial acceleration occurring at the remoteinertial platform 38, the accelerometers (not shown) within the inertialplatform 38 are essentially useless for high frequency motioncompensation purposes. Thus, the head mounted IMU 24 is utilizedprimarily for compensation of motion of the antenna 14, while theinertial platform 38 is used for navigation and as a vertical referencefor gravity compensation of the IMU 24, although both units mutually aideach other, via a coordinate transformation process to be described indetail hereinbelow.

The uncompensated motion of the antenna 14 in a given radial directionmust be constrained to satisfy the requirements imposed by the variousmapping and tracking modes. The constituent sources of error in anyestimate of the radial acceleration fall into three categories: (1) themagnitude of the velocity error; (2) the uncertainty in the angularorientation of the head coordinates with respect to the coordinatesinstrumented by the inertial platform 38 for the purpose of gravitycompensation; and (3), the error in acceleration measured by eachaccelerometer along its input axis. For map focusing purposes, theangular uncertainty between local vertical (as measured by the inertialplatform 38) and the input axis of the boresight or radial accelerometer171 (FIG. 8B) that must be held within a specified tolerance.

As mentioned above, the herein contemplated motion compensation systememploys head mounted gravity compensated accelerometers (two of whichare not shown but are within the IMU 24). The requisite gravitycompensation is a two part process. First, the change with altitude ofthe magnitude of gravity must be estimated, and secondly, the componentof gravity along the input axes of the accelerometers must becalculated. The variation of the magnitude of gravity relative to someground reference is given by:

    Δg=-2g.sub.e H/R.sub.e                               Eq. (2)

where g_(e) is the magnitude of the gravity vector at the groundreference point, H is the altitude of the aircraft 302, and R_(e) is theradius of the earth at the ground reference point. The second part ofcompensation for gravity is somewhat more complex, requiring a specialversion of the standard acceleration matching technique for determiningthe direction cosine matrix between the inertial platform 38 and thecoordinates of the accelerometers. The IMU 24 and the inertial platform38 then are used in combination, via the matrix of direction cosinesdefined by the coordinate axes of each instrument, to resolve thecomponents of gravity as required.

Before proceeding, it will be instructive at this point to brieflydescribe the antenna gimbal structure 200. Basically, the antenna gimbalstructure 200 allows the beam to be transmitted along the requisiteline-of-sight and isolates the antenna 14 from the motions of theaircraft 302. The antenna gimbal structure 200 comprises a four-axisgimbal arranged to decouple and stabilize the antenna 14 in the presenceof aircraft motion (roll, pitch and yaw). The contemplated configurationfurther allows all of the cardinal axes to pass through a common point,thereby permitting servo control about all the axes by means of the headmounted IMU 24. It should be noted here that, as mentioned above, afour-axis gimbal assembly is provided to prevent a gimbal lock situationwhich otherwise might occur with a conventional three-axis gimbalsystem. Thus, for example, a conventional three-axis gimbal system wouldexperience yaw gimbal lock if the antenna 14 were looking forward anddown 45° and the aircraft 302 were to pitch upward by 45°. Furthermore,even in the absence of gimbal lock, servo performance degrades as thepointing angle approaches within 30° of gimbal lock, thereby comprisingthe requisite pointing accuracy.

Given that a four-axis gimbal structure 200 is required to meet thetarget tracking requirements, a gimbal order of outer roll, pitch,azimuth and elevation waas selected as the optimum configuration fromthe point of view of, inter alia, stiffness, damping of resonances, easeof implementing the waveguide connection to the antenna 14 and gimbaltravel requirements. In such a configuration, the roll axis 211 and thepitch axis 203 form a stable platform for a conventionalazimuth/elevation mount. The yaw axis 201 and the elevation axis 209then form the part of gimbal structure 200 required in the targettracking modes.

Referring now also to FIG. 8B, the roll gimbal 213 and the pitch gimbal215 are stabilized by means of rate gyros 217, 219, respectively, whichare mounted on their respective axes. The antenna 14 is rate stabilizedabout the yaw (azimuth) gimbal axis 201 and the elevation gimbal axis209 by means of an azimuth rate gyro 221 and an elevation rate gyro 222.The roll gimbal 213 and the pitch gimbal 215 effectively isolate theazimuth gimbal axis 201 and the elevation gimbal axis 209 from thedynamic environment of the aircraft 302.

The head mounted IMU 24, which, as explained, is provided forcompensation of motion of the antenna 14 relative to the aircraft 302,comprises three mutually orthogonal rate gyros, the radial accelerometer171 and a pair of cross accelerometers, such elements (with theexception of the radial accelerometer 171) being mounted on theelevation gimbal 223 to serve also as a counterweight to the antenna 14.

The roll gimbal 213, the pitch gimbal 215, the elevation gimbal 223 andthe yaw gimbal 224 are driven by conventional torque motors of whichonly the roll torquers 225 and the pitch torquers 226 are shown. Suchmotors here are direct current motors with direct drive to avoid thebacklash associated with the use of gear drives.

The roll and pitch axes 211, 203, respectively, are rate controlled byoperation of a roll rate gyro 217 and the pitch rate gyro 219 inconjunction with rate control commands received from the digital dataprocessor 30 (FIG. 1). The inner gimbal axes, the yaw gimbal axis 201and the elevation gimbal axis 209 are all controlled by means ofcoordinate transformation resolvers (not shown) mounted on theirrespective gimbals. The rate integrating gyros (not shown) within theIMU 24 are used to sense rotation about the yaw gimbal axis 201 and theelevation gimbal axis 209. The output signals from the former are fedback to both the coordinate transformation resolvers (not shown) and thedigital data processor 30 (FIG. 1). It should now be appreciated bythose of skill in the art that the coordinate transformation resolvers(not shown) are utilized to control the yaw and elevation gimbal axes201, 209, respectively, to effect a higher control rate and therebyprevent undesired resonances. That is to say, the outer roll and pitchgimbal axes 211, 203, respectively, are controlled by feeding datadirectly back from their respective rate gyros 217, 219 to the digitaldata processor 30 (FIG. 1) at a relatively low data rate, here 64 Hz.Such a data rate is, however, too low to provide the requisite degree ofrate control of the yaw and elevation gimbal axes 201, 209,respectively, and, therefore, the coordinate transformation resolvers(not shown) are provided to effect a higher data rate while maintainingan identical interface with the digital data processor 30 (FIG. 1).

It should be noted here that the antenna gimbal structure 200 is uniqueand provides the following advantages vis-a-vis conventional two orthree axis gimbal structures: (a) the four axis gimbal provides thegreatest angle coverage for the antenna 14; (b) it reduces the dynamicenvironment seen by the yaw axis 201 and the elevation axis 209, therebyachieving the requisite stabilization; (c) it permits placing the yawaxis 201 perpendicular to the slant plane; and (d) the four axis gimbalstructure 200 allows target tracking without rotation of the antenna 14,thereby providing a signal processing advantage by easing cluttercancellation requirements. Furthermore, the inverted roll gimbal 211design permits the placement of the head mounted IMU 24 and acounterweight (not numbered) directly in line behind the antenna 14,making the horizontal and vertical balance about the pitch gimbal axis203 a symmetrical balance operation. This approach serves to produce thelightest possible system weight for a given antenna size.

Transfer Alignment (FIG. 8B)

As was mentioned above, the herein contemplated coordinatetransformation technique involves resolving the direction cosine matrixbetween the instrumented axes of the inertial platform 38 and the headmounted IMU 24. To this end the strapdown processor 173 (FIG. 7C)(sometimes hereinafter referred to simply as processor 173 and which ishere a Texas Instruments Model 9900 microprocessor) is shown to receivethe output signals from the head mounted IMU 24 and from a line-of-sightsensor assembly (not numbered) mounted at the rear center of the antenna14. The line-of-sight sensor assembly (not numbered) comprises theradial accelerometer 171 and the elevation rate gyro 222, both of whichare aligned with the antenna boresight axis 205. The processor 173 isphysically located within a boresight sensor assembly 36 (FIG. 1). Itshould be noted here in passing that the output signals from the IMU 24comprise the output signals from three mutually orthogonal rate gyros(not shown), a pair of cross accelerometers (also not shown) and theline-of-sight sensor assembly (not numbered). It should also be notedthat the process of transfer alignment in inertial navigation systems iswell known to those of skill in the art (e.g. "The Kalman Filter inTransfer Alignment of Airborne Inertial Guidance Systems" by A.Sutherland and A. Geib, TASC Report TR-134-2, Jan. 15, 1968; "StrapdownInertial System Alignment Using Statistical Filters: A SimplifiedFormulation" by J. Deyst and A. Sutherland, AIAA Journal, April 1973,pp. 452-6; and "A New Concept in Strapdown Inertial Navigation" by J. E.Bortz, NASA Technical Report R-329, March 1970) and will therefore notbe recounted in detail here. Suffice it to say that within the processor173 the signals from the boresight or radial accelerometer 171 and theaccelerometers (not shown) within the IMU are rotated, via a directioncosine matrix, into local vertical coordinates and integrated to obtainhorizontal velocity signals. High frequency changes in the directioncosine matrix are provided within the processor 173 by means of theinput signals from the three orthogonal rate gyros which sense theangular motion of the antenna axes. The horizontal velocity signals arecorrected for errors due to the displacement vector changes between theantenna 14 and the inertial platform 38 by being modified in a so-called"lever arm correction process" by the output signals from three rategyros (not shown) within the boresight sensor assembly 36 which isdisposed on the antenna bulkhead (not shown). The actual lever armcorrections are performed within the digital data processor 30 (FIG. 1).The thus corrected velocity signals from the digital data processor 30are passed to a microcomputer (not shown) associated with the inertialplatform 38 wherein they are compared, in a Kalman filter process, withthe horizontal velocity outputs obtained from the inertial platform 38.The difference in the velocities as determined by the inertial platform38 and the processor 173 are due to vertical and azimuth errors in thedirection cosine matrix as well as errors in the accelerometer and gyroinput signals to the processor 173. The Kalman filter process estimatesthe errors of the processor 173 and supplies these estimates to thelatter as correction signals.

Antenna Phase Center Motion Compensation (FIG. 8B)

Completing the contemplated motion compensation system are a pluralityof piezoelectric accelerometers (not shown, but which are here Model 606devices from Columbia Research Laboratory, Woodlin, Pa.) which areprovided to compensate for the effects of aircraft-induced vibrations onthe antenna 14. Thus, although the signals out of the line-of-sightsensor assembly (not numbered) may be used to compensate for the effectsof translational vibration, such signals are not useful in compensationfor rotational or differential (between quadrants) vibrations.Compensation for the effects of differential antenna vibration isespecially important in the ground moving target identification modewherein vibration-induced sidebands can appear as moving targets andthus seriously affect the reliability of the moving target detectionprocess. In such a mode the differential vibration of the antenna 14will phase modulate the radar return signals and cause sidebands (orfalse targets) to appear, thereby resulting in ambiguous Doppler anglemeasurements. Further, in the target tracking mode specular sidebandscan appear in the Doppler cell of a target being tracked, therebyproducing significant angle tracking errors. In addition, in the targettracking mode the degree of clutter cancellation between the antennaphase centers is directly affected by differential vibratory motionwhich places an upper limit on the cancellation obtainable, thereby alsodegrading angle tracking performance.

As was mentioned above, the phase centers of the antenna 14 aredisplaced with respect to each other and can rotate about axes in theplane of the antenna 14 passing through its geometric center. If thevibration-induced phase center displacements are not symmetric forcorresponding phase centers, the sum (Σ) and and difference (Δ) signalsderived will contain uncompensated vibratory phase modulation eventhough a radial accelerometer has been provided to compensate for thepure translatory motion.

Piezoelectric accelerometers (not shown) are used to measureaircraft-induced vibrations of the antenna 14. A total of five suchdevices are provided, one at each of the four antenna phase centers anda fifth located at the center of the antenna 14. The radialaccelerometer 171 is disposed adjacent to the piezoelectricaccelerometer (not shown) located at the center of the antenna 14. Itshould be noted here that the radial accelerometer 171 serves as areference for the piezoelectric accelerometers (not shown) which differfrom conventional accelerometers in that their frequency range is from 2Hz to 40 KHz.

It should now be appreciated that since the piezoelectric accelerometers(not shown) are high frequency devices, they do not measure the lowfrequency (linear or rotational) components of the inertialaccelerations at the antenna phase centers and the geometric center ofthe antenna 14. It can therefore be shown that, assuming a linear chirpwaveform is transmitted, the compensation signal for a given subaperturephase center would be of the form ##EQU2## where y_(i) is thedifferential displacement estimate, which is given by ##EQU3##

The P_(im) term in equation (4) corresponds to the piezoelectricaccelerometer signal from each phase center and the P_(am) term is thesignal from the piezoelectric accelerometer disposed at the center ofthe antenna 14. Thus, the signals from the piezoelectric accelerometers(not shown) disposed at the antenna phase centers are differenced withthat from the piezoelectric accelerometer disposed at the geometriccenter of the antenna 14, and the resultant signal is integrated, inanalog fashion, over the coherent integration time, T. It should benoted here in passing that the requisite integration is performed in theI.F. receiver 20 (FIG. 6) wherein compensation signals are developed tocontrol phase shifters in respective channels of such receiver. Itshould also be noted here that the actual displacements given byequation (4) are formulated each PRI. These displacement values aresubsequently modified by four different constants, corresponding towavelength values for each of the four frequencies utilized during agiven dwell, to develop the control signals for the phase shifters 351L,351U (FIG. 6).

Radar Synchronizer (FIGS. 9A-9F)

Before proceeding with a detailed description of the radar synchronizer34 (FIG. 1), a brief discussion of its function and design requirementsis in order. Thus, referring briefly back now to FIG. 1, the primaryfunction of the synchronizer 34 is real time control of the radar system10. As such, data and control commands from the digital data processor30 (FIG. 1) in the form of digital words are interpreted by thesynchronizer 34 and translated into timing and control signals foroperating such units as the I.F. receiver 20 (FIG. 1) and the exciter 32(FIG. 1). Among the key timing functions provided by the synchronizer 34are waveform gate generation for the exciter 32, analog-to-digital (A/D)sampling strobes for the I.F. receiver 20 and mode control switching forthe power distribution network 16.

Another function of the synchronizer 34 is high speed calculation. Thatis to say, the radar system timing depends upon controlling the pulserepetition interval of the radar system 10 in order to maintain equalangular increments between the pulse transmission paths to the mapreference point as the aircraft 302 (FIG. 10) flies along during amapping action thereby to provide the optimum focal width for thegenerated maps, as will be explained in greater detail hereinbelow.Suffice it to say here that the forementioned angle equality must bemaintained in the presence of aircraft acceleration of up to 4 g's inmagnitude in any direction and requires that the instantaneous range tothe map reference point or target be known. Range and range rate aresupplied to the synchronizer 34 by the signal processor 26 (FIG. 1)every 7.7 m sec. However, a range integration at a rate much higher thanthe digital data processor update rate is required for smoothinterpolation within the synchronizer 34. The PRI variation computationsalso require cross-acceleration data which is provided by the strapdownprocessor 173 (FIG. 7C) from cross-acceleration measurements made at theantenna 14, as was explained hereinabove.

Within the synchronizer 34 the time spent performing calculations aswell as the basic timing resolution and accuracy are determined by therequisite map resolution and tracking accuracy. Thus, for example, inthe short pulse tracking mode the quantization noise is specified to beconsistent with a 66 dB receiver dynamic range. In consequence, then,the A/D strobes sent by the synchronizer 34 to the signal processor 26(FIG. 1) must be positioned consistently, with no more than thirteenpicoseconds (ps) root mean square (rms) jitter with respect to thetransmitted waveform, to achieve this dynamic range.

The actual timing of the A/D strobes for the signal processor 26 isdependent on the range to the target or map reference point. Theforegoing jitter constraint imposes a parallel requirement on how oftenthe range word must be updated, for the reason that smooth transitionsshould occur in the range word to achieve stable mapping and tracking.The requisite range granularity considered together with the maximumspecified aircraft velocities here define an update rate of at least 40KHz for the range word.

As mentioned hereinabove, the digital data processor 30 (FIG. 1)calculates and delivers to the synchronizer 34 the range, R_(o), andrange rate, R_(o), every 7.7 msec. The necessity for instantaneous rangeevery 25 μs means that a range calculation must be made within thesynchronizer 34 at a 40 KHz rate. The range calculation is defined bythe following equation: ##EQU4## where: R_(EST) is the instantaneousestimated range;

R_(o) is the range calculated in the digital data processor 30 (FIG. 1);

R_(o) is the range rate calculated in the digital data processor 30(FIG. 1);

R_(o) is the radial acceleration calculated in the strapdown processor173 (FIG. 7C); and

Δt equals 25 microseconds.

The range calculation is reset and restarted in synchronism with theradar dwell rate timing as soon after a signal processor update aspossible.

The other major computational requirement is PRI variation. The PRIcontrol processing within the synchronizer 34 will be described indetail hereinbelow. Suffice it to say here that the basic or minimum PRIis set by the digital data processor 30 (FIG. 1) for each dwell or groupof dwells by sending a distance increment, Δd, to the synchronizer 34 ata 128 Hz rate. Every 25 μs the strapdown processor 173 (FIG. 7C) sendsto the synchronizer 34 a velocity word based on accelerometermeasurements. The synchronizer must predict the next velocity using a GHfading memory predictive filter and interpolate "instantaneous"velocities in real time. The rapidly updated (100 KHz) velocity isintegrated by the synchronizer 34 at a 100 KHz rate to develop a travelword, X.sub.⊥M. During a dwell, an accumulating sum of Δd's (i.e. thedesired amount of travel, X.sub.⊥D) is compared to X.sub.⊥M. WhenX.sub.⊥M equals X.sub.⊥D, a PRI timing sequence is initiated. At thesame time, another Δd is added to the previously accumulated X.sub.⊥D.The process repeats when X.sub.⊥M reaches the new X.sub.⊥D.

The expression for X.sub.⊥M is given by: ##EQU5## It should now beappreciated that the terms on the right hand side of Eq. (6) are the"MOTION COMPENSATION" signals of FIG. 7C. The synchronizer 34 then isoperative to compare X.sub.⊥M and X.sub.⊥D every 10 microseconds and tocommand a pulse to be transmitted whenever equality occurs. Sufficientaccuracy in PRI variation is achieved if X.sub.⊥M is calculated at a 100KHz rate. After each calculation X.sub.⊥M is compared to X.sub.⊥D todetermine if the quantities are equal. A time granularity of 10 μs isadequate for satisfaction of the clutter nulling requirement.

Digressing briefly here now for a moment and referring to FIGS. 12A and12B, the effect of PRI control is illustrated. Thus, during a givenmapping interval starting at time t_(o) and proceeding to time t_(n),the flight path of the aircraft 302 (FIG. 7A) is shown to graduallyapproach the map center point, O. It is apparent that between pulsetransmission times t_(o) and t₁ the aircraft 302 (FIG. 10) must traversea greater distance than that required between pulse transmission timest_(n-1) and t_(n). It should also be apparent that the angularincrement, δθ, swept out between pulse transmission times t_(o) and t₁,is identical to that swept out between times t_(n-1) and t_(n). Thus, itcan be seen that the effect of PRI control is to insure that, despitethe motion of the aircraft 302 (FIG. 10) equal angle increments areswept out between the pulse transmission times. For the flight pathillustrated the PRI will decrease exponentially during the dwell period.

As mentioned above, a PRI variation is needed to attain a satisfactorywidth of a generated map when the aircraft 302 is maneuvered during adwell. A greater depth-of-focus of a generated map is realized byvarying the A/D strobe frequency slightly during each dwell as afunction of change in the depression angle, C_(o) (FIG. 7A). As the A/Dstrobe frequency need only be changed at relatively infrequentintervals, the 128 Hz update rate of the synchronizer 34 by the digitaldata processor 30 is adequate for satisfactory depth-of-focus and thereis therefore no requirement for real-time calculation in thesynchronizer 34 to perform this function.

Referring briefly now to both FIGS. 1 and 9A, the radar synchronizer 34is shown to include a data input/output (I/O) controller 371, amicrocomputer 375, a timing and control logic unit 377, a clockgenerator 379, and a 100 MHz coherent oscillator (COHO) 381. The designof the radar synchronizer 34 is driven by the requirement to communicatewith several subsystems other than the digital data processor 30. Thus,for example, the digital data processor 30 sends, either at a 128 Hzrate or once per radar dwell, approximately 256 digital words (in 16 bitserial form) to the radar synchronizer 34 and the strapdown processor173 (FIG. 7C) delivers a digital word (18 bits) every 25 μs. Further,before every dwell the radar synchronizer 34 must reprogram the exciter32 (FIG. 1) with a 96-bit serial word, and the transmitter 22 requiresan 8-bit word for power level control. Still further, before every PRIthe signal processor 26 requires a PRI normalization factor which is thelength of the previous PRI. The data links which are provided by thedata I/O controller 371 are, therefore, data outputs to the exciter 32,the transmitter 22, the R.F. receiver 18, the I.F. receiver 20 and thepower distribution network 16 and a data input from the strapdownprocessor 173 (FIG. 7C) and a bi-directional data link to the digitaldata processor 30 and the signal processor 26.

Referring now to FIG. 9B, the data I/O controller 371 is shown toinclude a data I/O module 373, a microprocessor 383 and a shared memorymodule 385. The data I/O module 373, which here may be comprised of aplurality of shift registers (not shown), provides the interface betweenthe radar synchronizer 34 and the other major systems of the radarsystem 10 (FIG. 1). An exemplary data interface, here that with thedigital data processor 30 (FIG. 1), is shown to include both data inputand output lines. The data output lines include a request output (REQO)line, a data output (DATO) line, a clock output (CLKO) line, and anacknowledge (ACKO) line. The data input lines include a request input(REQI) line, a data input (DATI) line, a clock input (CLKI) line and anacknowledge (ACKI) line. A data output transfer from the data I/O module373 is initiated by changing the REQO line from a logic 0 level to alogic 1 level. If the digital data processor 30 is ready to accept datathe ACKI line is switched from a logic 0 to a logic 1 level. The dataI/O controller 371 then gates ON (changes to a logic 1 level) the CLKOline and outputs, synchronously with the clock signals, a stream of dataover the DATO line. The falling edge of the first clock pulse causes thedigital data processor 30 to reset its ACKI output signal back to alogic 0 level. Data transfer from the digital data processor 30 to thedata I/O module 373 is similarly accomplished with the digital dataprocessor 30 initiating the REQ signals for data input and outputtransfers.

Within the radar synchronizer 34 all the high speed time-criticalcomputations are performed within the microcomputer 375. The data I/Ocontroller 371 acts as a buffer between the microcomputer 375 and thevarious subsystems of the radar system 10 (FIG. 1) to prevent interruptsof such microcomputer when data transfers are required. Such bufferingis necessary because the microcomputer 375 may have to perform severalconcurrent calculations at both a 40 KHz and 100 KHz rate, withconsistency in the timing of such calculations determining the accuracyof both the PRI variation and the A/D strobe frequency for the IFreceiver 20 (FIG. 1). It follows, then, that interrupts to themicrocomputer 375 are inconsistent with the task of timing the PRI.Backing up incoming data from the digital data processor 30 (FIG. 1) ina first-in, first-out memory and sampling it on a polled basis isfeasible only for small amounts of data. The only architecture generalenough to handle the requisite data transfers is one that features anintelligent data controller. Thus, the data I/O controller 371, althoughit is a peripheral of the microcomputer 375, contains local intelligenceprovided by the microprocessor 383, which is here a 2900 Seriesmicroprocessor from Advanced Micro Devices, Inc., Sunnyvale, Calif.,94086. The microprocessor 383 peels off synchronizer control words fromthe data stream provided by the digital data processor 30 (FIG. 1) andplaces them, together with an updated status word, in a smallscratch-pad memory compartment of the shared memory module 385. Themicrocomputer 375 thus has a small group of data words to process and isprotected from the necessity of directly servicing the radar subsystems.

The data I/O controller 371 is able to respond to data transfer requestson an interrupt basis without affecting radar timing since it processesdata in parallel. At initialization of the radar system 10 (FIG. 1), thedata I/O controller 371 takes priority over the microcomputer 375 byhalting the latter and performing direct memory address (DMA) datatransfers into the shared memory module 385 and the control memorymodule 387 within the microcomputer 375.

The shared memory module 385 comprises a direct memory address (DMA)logic section (not shown) and a dual port random access memory (RAM)(also not shown) which is shared by the microcomputer 375 and themicroprocessor 383. The dual port RAM serves as a macroprogram memoryfor the microcomputer 375 and, in addition, stores: (a) mode data forthe timing and control unit 377 (FIG. 9A); and (b) numerical data forthe microcomputer 375 and the timing and control logic unit 377 (FIG.9A). It should be noted here that all the data transfers within theradar synchronizer 34 are accomplished by data busses (not numbered)comprising exposed registers (exposed register module 395), which willbe described in detail hereinbelow.

It should be recalled here that all of the high speed, time-criticalalgorithms (as, for example, the PRI variation computations) areperformed within the microcomputer 375. The update rate and theprecision required for these calculations as well as the desire to avoidutilizing double precision arithmetic lead to the choice of a 24-bitbipolar bit-slice implementation for the microcomputer 375. Thebit-slice family chosen for implementing the latter is the 2900 seriesavailable from Advanced Micro Devices (AMD). Thus, the sequencer 389 ishere a 2910 series microprogram controller and the arithmetic logic unit(ALU) module 391 is comprised of six 4-bit 2903 microprocessor slices toachieve the requisite 24-bit machine width. It should be noted here thateach of such slices includes sixteen randomly accessible dual-portgeneral purpose internal registers with the necessary "hooks" to expandthe register file by combining external (off-chip) and internalregisters. The multiplier module 393 is here comprised of a plurality ofMPY24HJM multipliers from the LSI Products Div. of TRW, Redondo Beach,Calif. and the memory module 397 is here a random access memory becauseof the versatility of such a device.

Completing the microcomputer 375 is an exposed register module 395 whichhandles all the data transfers from the microcomputer 375 to the dataI/O controller 371 and the timing and control logic unit 377. Such unitis implemented using emitter-coupled logic (ECL) and, therefore, aTTL/ECL interface exists between the microcomputer 375 and the timingand control logic unit 377 (FIG. 9A). Because the length of each PRImust be controlled to within 10 μs, a very high premium is put on therapid execution of the PRI algorithm within the microcomputer 375. Partof that execution time is expended in outputting the PRI start commandto the timing and control logic unit 377 (FIG. 9A). Performinginput/output I/O instructions with any conventional parallel outputtechnique is too wasteful of the very limited execution time availableto the microcomputer 375 within a PRI. The exposed register module 395solves the data transfer problem by taking advantage of the registerexpansion capability within the module ALU 391. Thus, it should berecalled that each 2903 chip in the ALU module 391 contains sixteengeneral purpose internal registers as well as "hooks" permitting theconnection of additional off-chip registers. A plurality of suchoff-chip registers (not shown) which are assigned dedicated functionsare provided in the exposed register module 395. These off-chipregisters are controlled in the same fashion as the on-chip registers,and TTL to ECL level shifters (also not shown) present the contents ofthese off-chip registers directly and in parallel to the ECL timinglogic within the timing and control logic unit 377 (FIG. 9A). Thus,programming the ECL logic from the exposed register module 395 requiresno processing time overhead specifically for output operations. That isto say, as soon as a control word is calculated in the microcomputer375, control of the ECL logic within the timing and control unit 377(FIG. 9A) takes effect because the result of each calculation alwaysresides in the dedicated exposed register (not shown) within the exposedregister module 395. Eight control words, some of which change four tosix times per PRI, are handled in this fashion thereby savingsignificant amounts of processing time during each PRI.

Before proceeding with a detailed description of the timing and controllogic unit 377 it should be noted here that the timing of the radarsystem 10 (FIG. 1) is performed in two different sections of thesynchronizer 34. Thus, PRI variation and scheduling is performed withinthe microcomputer 375, while the timing within a PRI is determined byECL logic within the timing and control logic unit 377 which iscontrolled by, but is much faster than, the microcomputer 375. It shouldalso be noted that the logic circuitry within the timing and controllogic unit 377 is divided into two halves corresponding, respectively,to transmit and receive mode timing. The transmit and receive modecircuitry are similar, differing only in that the receive mode circuitrydoes not include either a phase lock loop module or a range countermodule because the receive mode clock is here generated in the phaselock loop module 401 (FIG. 9E). Thus, for the sake of drawing clarity,only that portion of the circuitry corresponding to the transmit modetiming will be shown and described in detail.

Referring now to FIGS. 9D to 9F, the transmit mode timing logic withinthe timing and control logic unit 377 is shown to include a "modetransparent, real time" control module 399, a phase lock loop module401, and a range counter module 403, all of which receive controlsignals from the microcomputer 375 via exposed registers (not shown)within either the exposed register module 395 (FIG. 9C) or themultiplier module 393 (FIG. 9C). The basic clock for the transmit modetiming is the 50 MHz master digital clock (MDC) which is obtained fromthe clock generator 379 (FIG. 9A), and which is developed within thelatter by dividing down the 100 MHz signal received from the 100 MHzCOHO 381 (FIG. 9A).

The mode transparent real time control module 399 is required togenerate, within a given PRI, control signals for the various radarsubsystems as, for example, spoof frequency control signals for theexciter 32 (FIG. 1), power level control signals for the transmitter 22(FIG. 1), and receiver gain levels for the R.F. receiver 18 (FIG. 1).Prior to the start of a given radar dwell (which could total 1,000PRI's) the microcomputer 375 (FIG. 9C) clears the memory address counter407 and then loads that counter (and the random access memory (RAM) 405)with real-time control states interleaved with state-change times. Atthe start of each PRI, the microcomputer 375 (FIG. 9C) resets a PRItimer 409, which is here a sixteen bit counter, with a wordcorresponding to the length of the desired PRI, which could range from100 μs to 1.3 ms. The memory address counter 407 and the PRI timer 409are then synchronously clocked by the 50 MHz MDC. The memory addresscounter 407 addresses the first event time-stored in the RAM 405(expressed as a sixteen bit binary number), which event is then clockedinto a next event time register 411. A comparator 413 detects when thePRI timer 409 reaches such number stored in the next event time register411. When such detection occurs, the comparator 413 provides a controlsignal to an event control mode logic unit 415, which is here ofconventional design, and which, in turn, causes the memory addresscounter 407 to be bumped twice, once to a clock a control word from theRAM 405 to an output register 417 and second time to clock a new eventtime word into the next event time register 411. It should be noted herethat as only changes of state (pulse edge scheduled times) are stored inthe RAM 405, each data line leaving the output register 417 is adiscrete control line.

Before proceeding with a detailed description of the phase lock loopmodule 401, the rationale for providing such a unit will be described.Thus, in order for the radar system 10 (FIG. 1) to reliably track atarget with the precision desired here, a time resolution of 140 ps in266 μs is required. Further, in the short pulse tracking mode thequantization noise is here specified to be consistent with a 66 dBdynamic range of a receiver. The A/D strobes sent by the synchronizer 34(FIG. 1) to the I.F. receiver 20 (FIG. 1) must, therefore, be positionedconsistently with no more than 13 ps RMS jitter (with respect to thetransmitted waveform) to achieve such a dynamic range. The timing of theA/D strobes for the I.F. receiver 20 (FIG. 1) depends on the range tothe target or map reference. The range and therefore the time resolution(140 ps) required relative to the maximum specified system range dictatethat the range be known to eighteen binary places. The jitterrequirement adds three more least significant bits (LSB's), bringing thetotal number of bits needed up to 21. Another result of the jitterconstraint is a parallel requirement on how often the range word must beupdated within the microcomputer 375 (FIG. 9C). Finally, from update toupdate smooth LSB transitions should occur in the range word toaccomplish mapping and tracking without danger of the occurrence ofexcessively large transients.

The foregoing considerations dictated the requirement that thetransmit-to-receive time interval be timed with 13 ps quantization and39 ps resolution. The direct digital implementation of a range counterhaving 39 ps time increments requires a 25.6 GHz clock frequency.Because no commercially available logic family supports clockfrequencies of that order of magnitude, a mixed analog and digitalapproach is used here to achieve the desired time resolution.

Thus, it will be appreciated by those of skill in the art that, ingeneral, by introducing an analog offset voltage into the control loopof a phase-locked voltage-controlled oscillator (VCO), the phase of theVCO can be moved away from that of the phase detector referencefrequency in a very controlled manner. Very precise departures in phasecan be achieved by using an accurate digital-to-analog (D/A) converterto generate the analog offset voltage. Thus, if the phase of a VCO wereto be locked to the 50 MHz MDC, which, as explained, controls thetransmit mode timing, the phase of such VCO could be controlled over a 0to 2π radian phase shift interval (corresponding to a 0 to 20 ns timeinterval). In consequence, then, a very fine time resolution can beachieved using a nine bit D/A converter to attain phase adjustments in0.70° increments (or 39 ps time increments). If a twelve bit D/A isutilized with the 3 LSB's grounded, time uncertainty (or jitter) iscontrolled down to less than 5 ps, thereby satisfying the 13 ps jitterspecification mentioned hereinabove.

It should now be appreciated that the phase-lock loop module 401 may beprovided to develop clock signals for receive mode timing. Thus, suchmodule is shown to include a first and second phase-lock loop (notnumbered) which develop, respectively, a phase slipped clock (PSC)signal and a frequency slipped clock (FSC) signal. The PSC is usedwithin the range counter module 403 (FIG. 9F) for the receive modetiming, while the FSC signal is used to provide the requisite A/D strobefrequency variation which, as mentioned hereinabove, is required tocontrol the depth-of-focus of the generated maps. The first phase lockloop (not numbered) comprises a digital phase detector 419, thereference signal to which is the 50 MHz MDC, a loop control amplifier421, which here may be an operational amplifier, a loop filter 423 and avoltage-controlled crystal oscillator VCXO 425. A second control signalis applied to the loop control amplifier 421, via a range register 427(a 24 bit device) and a converter 429 (a 12 bit device with the threeleast significant bits grounded). The range register 427 is, at thestart of a given dwell, loaded with a 24 bit estimated range word,R_(EST) (which is also a measure of the transmit-to-receive timing) andupdated by the microcomputer 375 (FIG. 9C) at a 40 KHz rate. The nineLSB's of the estimated range word, R_(EST), from the range register 427are translated to an analog voltage with 12 bit accuracy by the D/A 429wherein the 3 LSB's are grounded. Thus, nine LSB's of the estimatedrange word, R_(EST), are used to develop the PSC for the receive modetiming. The thus described phase lock loop (not numbered) is effectiveto divide one 20 ns 50 MHz clock cycle into 512 parts, each of 39 psduration. The nine LSB's of the estimated range word, R_(EST), set aphase delay for the PSC in terms of these 39 ps time units.

A practical problem arises when very little phase difference existsbetween the master digital clock (MDC) and the phase slipped clock(PSC). In this event, the range counter module 403 (FIG. 9F) could beenabled just before the rising edge of the PSC. The requisite setup timefor the range counter module 403 would not be met in this instance. Thisproblem is dealt with through the use of a comparator 439 to detectmarginal timing situations in the range word, R_(EST). The output of thecomparator 439 is combined in an exclusive OR gate 441 with the 50 MHzMDC signal. Reference numbers, corresponding to the ranges occurring 5ns before and after R_(EST) are stored in the comparator 439. When thenine LSB's of the range word, R_(EST), from the range register 427 fallbetween these reference numbers the comparator 439 causes the phase ofthe 50 MHz MDC out of the exclusive OR gate 441 to be inverted. Suchreversal, as will be explained in detail hereinbelow, ensures that therange counters within the range counter module 403 (FIG. 9F) are enabledwell in advance of the first PSC signal to which they must respond.

As mentioned hereinabove, an A/D strobe frequency variation must beincluded to provide for the depth of focus of the generated maps. ThisA/D strobe frequency control is provided by means of the second phaselock loop (not numbered), which here comprises a phase comparator 431,the reference signal to which is the PSC signal, a loop controlamplifier 433, which may be an operational amplifier used as a summingamplifier, a loop filter 435 and a VCXO 437. The output signal from thephase lock loop (not numbered) is the frequency slipped clock (FSC)signal which is passed ultimately to the I.F. receiver 20 (FIG. 1). Theloop control amplifier 433 is shown to receive a pair of controlsignals, a first one of which is received from a phase register 443 (12bits) via a D/A 445, (12 bits with the 3 LSB's grounded). The secondcontrol signal to the loop control amplifier 433 is received from arange register 447 (12 bits), via a D/A 449 (12 bits with the 3 LSB'sgrounded), and an integrator 451, comprising a switch S1, a capacitor C1and an amplifier A1. The microcomputer 375 (FIG. 9C) provides the phaseregister 443 with a phase offset word and the range register 447 with arange offset word. The phase offset word and the range offset words areeffective to produce the FSC having a phase ramp (frequency offset) withboth an adjustable starting phase and phase slope which are dependent onmeasured, real-time motion compensation parameters provided to themicrocomputer 375 (FIG. 9C). It should now be appreciated by those ofskill in the art that the FSC signal is a coherent transform of the PSCsignal and that the starting phase and phase slope of the FSC signal arereal time functions of aircraft motion. Finally, it should be noted thatthe integrator enable signal to the integrator 451 is provided a signalfrom the timing and control logic unit for the receiver (which unit hasnot been illustrated but is, as noted above, similar to the timing andcontrol logic unit being discussed).

Referring now to FIG. 9F, the range counter module 403 is shown toinclude four J-K flip-flops (f/f) 453₁, 453₂, 453₃ and 453₄, each ofwhich controls a corresponding counter 455₁, 455₂, 455₃ and 455₄ to makeup four counter pairs (not numbered). It should be noted here that fourcounter pairs are required to control the four transmitted frequenciesand that a different number of such pairs would be used if the number ofpossible transmitted frequencies were changed. The R.F. receiver 18(FIG. 1) and the I.F. receiver 20 (FIG. 1) are activated and strobed bythe synchronizer 34 (FIG. 1) to pass the desired return signal from anytransmitted pulse, even though such signal could occur at any time (evenafter the transmission of a third transmitted pulse). The requisiterange timing and gating must be done concurrently for all four pulsesbecause of the interleaved nature of system timing and, therefore, fourindependent range counters are required. Thus, the flip-flops 453₁ . . .453₄ are shown to be clocked by the 50 MHz noncoincidence MDC obtainedfrom the exclusive OR gate 441 within the phase lock loop module 401,while the counters 455₁ . . . 455₄, which are all sixteen bit devices,are clocked by the 50 MHz PSC. The data input to the J data terminals ofeach of the flip-flops 453₁ . . . 453₄ are provided by the transmitsequence control lines (not numbered) from the output register 417within the mode transparent real time control module 399 (FIG. 9D).

The J data terminals of the flip-flops 453₁ . . . 453₄, in conjunctionwith the Q output terminals, provide enabling signals to the counters455₁ . . . 455₄. The latter receive via an exposed register bus (notnumbered) the fifteen MSB's of the range word, R_(EST), from the rangeregister 427. Once enabled, each of the counters 455₁ . . . 455₄ beginto count the range word, R_(EST), and upon completion of the count, eachprovides an output signal at the logic 1 level. The output signals fromthe counters 455₁ . . . 455₄ are provided as input signals to the K dataterminals of their respective flip-flops. It will be appreciated bythose of skill in the art that when both the J and K terminals of theflip-flops 455₁ . . . 455₄ are at the logic 1 level and a clock pulse isreceived, the Q output will change state. Advantage is taken of thisproperty of the flip-flops 453₁ . . . 453₄ to provide enabling ordisabling signals to the counters 455₁ . . . 455₄.

Synchronously with the terminal count word provided to the flip-flops453₁ . . . 453₄, receive sequence start signals are sent, via an OR gate457, to the "receive" mode, real time control module (not shown, butcorresponding to that shown in FIG. 9D) to initiate the receive modetiming.

At this point in time a brief review of the PRI timing is in order. ThePRI is a cycle of four transmitted active pulses, each at a differentfrequency. Preceding the transmission of each active pulse is apanoramic analysis receive period and an additional transmitted pulse(spoof frequency). The spoof frequency pulses have unique frequenciesdifferent from that of the active pulse frequencies. The active pulsescould be either uncoded, frequency ramped, or phase coded. The timing ofthe just described features (i.e., the panoramic analysis, the spooffrequency transmissions and the active pulses) is synchronized to the 50MHz MDC.

Navigation and Position Update Modes (FIG. 10)

Before proceeding with a detailed description of the various operatingmodes of the radar system 10 (FIG. 1) together with the associatedsignal flow diagrams through the signal processor 26 (FIG. 1) and thedigital data processor 30 (FIG. 1), it will be beneficial at this pointto remember that, in addition to being used for weapon guidance, thatradar must also be used to solve the navigational problem involved. Partof such problem is that of navigating the aircraft 302 (FIG. 10) to aposition in space, relative to the target area, with sufficient accuracy(both in position and velocity) to make weapon delivery possible. Theproblem is compounded by the fact that the aircraft 302 (FIG. 10) mustoperate in a tactical environment, meaning that it will be exposed to ahostile ECM environment, adverse weather conditions and interdictingfire.

It will be recalled that the aircraft 302 (FIG. 10) is equipped with aninertial platform 38 (FIG. 1) which is utilized, inter alia, fornavigation of the aircraft 302 (FIG. 10). Because the inertial platform38 (FIG. 1) is subject to long term drift errors which effect thenavigational accuracy, it is periodically updated with radar derivedvelocity and position data. The aircraft 302 (FIG. 10), in thenavigation mode, may be operating at an altitude of from 500 to 30,000feet which imposes a rather severe dynamic range requirement, as will beexplained in detail hereinbelow. To minimize the risk of detection, theradar system 10 (FIG. 1) will be operated only periodically and then forthe minimum amount of time. Furthermore, while penetrating enemyterritory the aircraft 302 will operate at the minimum altitude and willrise (pop up) to a high altitude only when required. In consequence,then, the system of synthetic aperture measurements taken for positionand velocity updates must reflect a great degree of invariance to theeffects of altitude.

As was explained hereinabove, synthetic aperture radar systems requireprecise measurements of antenna acceleration for motion compensation ofthe radar return signals. In order to satisfy operational requirementsand mounting constraints within the aircraft 302 (FIG. 10), the antenna14 (FIG. 1) and the inertial platform 38 (FIG. 1) are physicallyseparated. This physical separation causes relative motion between theantenna 14 (FIG. 1) and the inertial platform 38 (FIG. 1) due both toaircraft maneuvers and to the flexure of the airframe. As was alsoexplained hereinabove, the motion of the antenna 14 (FIG. 1) can bemeasured and therefore compensated by the high resolution inertialmeasuring unit 24 (FIG. 1), which is mounted in close proximity to theformer. The long term alignment of the latter is maintained by slavingit to the inertial platform 38 (FIG. 1).

Referring briefly back now to FIGS. 1, 8A and 8B, the method of transferalignment will be reviewed. Thus, the boresight sensor assembly 36,which comprises three orthogonal rate gyros (not shown), is mounted nearthe base of the antenna 14. The rate gyros (not shown) are aligned tothe aircraft roll, pitch and yaw axes 211, 203, 201, respectively. Thesensed angular rates of such gyros, together with the orientation andposition of the antenna 14 relative to the inertial platform 38 (asmeasured by the IMU 24), are used to determine the north and eastvelocity differences between such antenna and platform during aircraftdynamic maneuvers. The IMU 24 is mounted on the elevation gimbal 223(FIG. 8B) and comprises two accelerometers (not shown) and threeorthogonal rate gyros (also not shown). The third accelerometer requiredto complete the IMU 24 triad is the radial accelerometer 171 which ismounted at the rear center of the antenna 14 (FIG. 1). The roll rategyro (not shown) is aligned with the antenna line-of-sight (LOS) vector(not numbered), while the pitch rate gyro (not shown) and the Y-axisaccelerometer (also not shown) and the Z-axis accelerometer (also notshown) are mounted orthogonal to the roll and pitch rate gyros (notshown). The rate gyro outputs and the accelerometer outputs from the IMU24, as well as the accelerometer signal output from the radialaccelerometer 171 (FIG. 8B) are passed to the strapdown processor 173(FIG. 7C) wherein processing is carried out to generate earth-referenced(north, east and local vertical) mathematical gimbal axes as well asnorth and east velocities. Such velocities are compared within a Kalmanfilter (not shown but within the inertial platform 38) with inertialvelocities obtained from the inertial platform 38 to determine bothmisalignment angles and sensor errors within the head mounted IMU 24.These errors are then used to correct the output signals from the headmounted IMU 24. Antenna axis accelerations are sent from the strapdownprocessor 173 to the digital data processor 30 for motion compensationpurposes. This acceleration data corresponds to the output signals fromthe head mounted IMU 24 which have been gravity compensated as well ascorrected for bias and scale factor errors.

Referring now to FIG. 10, the aircraft 302 is shown to be flying along apath such that its velocity vector, V, is coincident with Y_(i) axis ofthe aircraft inertial reference frame X_(i), Y_(i), Z_(i). In thevelocity update mode, the radar system 10 (FIG. 1) illuminatessuccessive portions of the ground terrain (not shown) and processes thereturn data from at least three of such portions to derive the threecomponents of aircraft velocity. The theory of operation of Dopplerradar navigators is well known to those of skill in the art and willtherefore not be recounted. Suffice it to say here that the radar system10 (FIG. 1) may be thought of as transmitting a cone of constant Dopplerfrequency centered about the aircraft velocity vector, V. The cone anglecorresponds to the beam squint angle, A, which is the angle between theestimated position of the velocity vector, V, and the antenna boresightaxis 205 (FIG. 8B). The azimuth angle, B, is defined as the anglebetween the horizontal projections of the antenna boresight axis 205 andthe velocity vector, V. The elevation angle, C, is the angle of theantenna boresight axis 205 with respect to the horizontal groundterrain.

The intersection of a constant Doppler cone (not shown) with a terrainproduces a contour line of constant Doppler frequency referred to as anisodop 501, which is here in the form of a hyperbola when such terrainis horizontal. The particular shape of the isodop 501 is dependent onthe squint angle, A, and the height of the aircraft 302 above theterrain. If, therefore, the altitude of the aircraft 302 and the beamsquint angle, A, are held constant as the antenna 14 (FIG. 1) is movedfrom dwell to dwell, the antenna boresight axis 205 will be forced totravel along the isodop 501. It can be shown that the squint angle, A,the azimuth angle, B, and the elevation angle, C, are related asfollows:

    cos A=cos B cos C                                          Eq. (7)

Holding the squint angle, A, constant as the antenna 14 (FIG. 1) isscanned will therefore require changes in both the azimuth angle, B, andthe elevation angle, C. As a given squint angle, A, and aircraftaltitude will define a particular hyperbola, the requisite gimbalcontrol commands to cause the antenna boresight axis 205 to follow theisodop 501 may be computed from an a priori knowledge of the equation ofthe hyperbola that corresponds to the isodop 501.

It should be recalled here that the widest possible focal width mapsunder the conditions of aircraft acceleration are obtained bycontrolling the PRI of the radar system 10 (FIG. 1). It can be shown,however, that the PRI is also dependent on the squint angle, A. That is,as the antenna 14 (FIG. 1) is squinted from broadside toward the nose ofthe aircraft 302, the range to the illuminated terrain continuallyincreases. Thus, in order to avoid range ambiguity problems and toprevent pulse eclipsing, the PRF of the radar system 10 must decrease asthe antenna 14 (FIG. 1) is squinted toward the nose of the aircraft 302.Decreasing the PRF will result in a concomitant increase in the PRI. ThePRI control problem is further aggravated by the fact that the depth,width and height of focus of the illuminated terrain are all dependenton the cell resolution width, Δa, which, in turn, is dependent, interalia, on the height of the aircraft 302 above the terrain.

Under PRI variation the depth, height and width of focus are given,respectively, by ##EQU6## where Δa=(λ/2) (h/d) (n/N) (1/sin C) andn=number of dwells. Using the criteria given by the foregoing equationsensures that the azimuth cell resolution is sufficiently large for depthof focus to equal the transmitted beamwidth. Thus, as the antenna 14(FIG. 1) is slued to successive points on the isodop 501, the number ofpulses, N, for each successive dwell is computed from Equation (12).##EQU7##

As was mentioned above, the radar system 10 (FIG. 1) is equipped withboth an azimuth and elevation monopulse capability. Thus, range andrange rate (Doppler) measurements can be associated with a specificdirection in space. In order to use this capability to estimate thevelocity of the aircraft 302 the estimate of aircraft velocity obtainedfrom the inertial platform 38 (FIG. 1) is used to first estimate thecomponent of velocity along the antenna boresight axis 205. As wasmentioned hereinabove, the velocity data estimates from the inertialplatform 38 (FIG. 1) are compensated by data from the boresight sensorassembly 36 (FIG. 1) and are passed to the digital data processor 30(FIG. 1) wherein they are used in conjunction with estimates of thedirection cosine matrix obtained from the head mounted IMU 24 (FIG. 1)to develop estimates of the range and range rate to the point on theterrain in the antenna boresight direction. Radar measurements aboutthese estimated values of range and range rate are then made and theresulting data is used to determine the error in the range rate(Doppler) estimate.

Each beam or dwell measurement consists of data derived from each offour illumination frequencies. It is noted here in passing that thenumber of pulses for each of the four frequencies is determined fromEquation (12) prior to the start of each dwell. The radar return datafrom each antenna phase center (quadrant) and for each illuminationfrequency are passed through the R.F. receiver 18 (FIG. 1) and the I.F.receiver 20 (FIG. 1) wherein they are downconverted to in-phase (I) andquadrature-phase (Q) video signals. The I and Q signals are ultimatelydigitized and passed to the signal processor 26 (FIG. 1). Within thelatter the data samples are motion compensated in a manner to bedescribed in detail hereinbelow. The resulting motion compensated datasamples are pulse compressed and processed by means of a fast Fouriertransform (FFT) algorithm ultimately to form range-Doppler data arrays(N range cells by M Doppler cells) for the four antenna phase centersand for each illumination frequency. It should be noted here that forthe purpose of this discussion it is presumed that the transmittedpulses are binary phase encoded although, depending on the mappingrange, they could just as well be chirped or uncoded. The data from eachof the range-Doppler arrays are processed, in a known manner such as isdescribed in U.S. patent application Ser. No. 47,957 filed Jun. 11, 1979by Ogar et al and assigned to the assignee of the present application,to generate normalized azimuth and elevation monopulse data on acell-by-cell basis for the N by M array for each illumination frequency.The normalized azimuth and elevation monopulse data arrays areprocessed, in a manner described in the just recited application, todetermine the Doppler frequency associated with the zero-crossing pointof the monopulse discriminator curve which corresponds to the center ofthe radar beam. The difference between the predicted and measuredvelocity, as determined from the Doppler frequency associated with thezero crossing point, is the velocity error for the beam data beingprocessed.

The just described process is repeated for each beam and the resultingvelocity errors are combined with their associated beam direction cosinematrix to produce the components of velocity error in inertial referencecoordinates for updating the inertial platform 38 (FIG. 1).

POSITION UPDATE MODE (FIG. 10)

As mentioned previously, the inertial platform 38 (FIG. 1) isperiodically updated in order to maintain the long term position errorsso that the aircraft 302 (FIG. 10) can be navigated to a point (saypoint p, FIG. 10) in space under varying environmental conditions andaircraft dynamics with, here, an error of no more than 0.8 nauticalmiles. The position update mode entails mapping the terrain surroundinga predetermined checkpoint whose position is known and then determiningthe position errors of the inertial platform 38 (FIG. 1) by measuringthe difference between the position of such check point and the positionof the center of the radar beam in the map.

The position update mode initiated by the radar operator, in practice,at successive fifteen minute intervals during flight of the aircraft 302(FIG. 10). In the position update mode transmitter power is radiatedfrom a single antenna segment (say segments 14AS, FIG. 3A) in order toproduce a relatively broad antenna beam, thereby to illuminate as largean area as possible around the predetermined checkpoint. Return signalsare, however, processed in a manner similar to that describedhereinabove for the velocity update mode. That is, the map center whichcorresponds to the antenna boresight axis 205 is computed both inazimuth and elevation. The position update mode, the map produced byprocessing return signals is displayed to the radar operator. Using acursor (not shown) the operator designates the predetermined checkpointon the map. The offset of predetermined checkpoint with respect to themap center is then computed in antenna reference frame coordinates. Thecursor address of the predetermined checkpoint in range/Dopplercoordinates provides the basis for determining an offset error for suchcheckpoint with respect to the center of the beam. That offset error, inantenna space coordinates, is then combined in a known manner with thebeam direction cosine matrix to provide the components of position errorin inertial reference frame coordinates for updating the inertialplatform 38 (FIG. 1).

FIXED TARGET DETECTION (FIG. 11)

The radar system 10 (FIG. 1) is required to automatically detectstrackable targets (meaning targets of significant military values as,for example, armored columns, air craft hangars, runways, bridges, fuelsupply depots or air defense radars), both fixed and moving in thepresence of ground clutter, large point targets and EMC. The detectionof fixed trackable targets requires a screening process to sort the mapdata while the detection of moving targets and their relocation to theproper map position requires discrimination techniques between targetsand ground clutter. It should be noted here that the trackable targetconcept is important in that it determines the requisite map resolutionsize which, in turn, determines whether or not there will be a rangeslip problem during any dwell. That is to say, the amount of range slipduring a given dwell may be expressed as:

    Range Slip=(N.sub.RG /2) (λ/2ΔX)              Eq. (14)

where N_(RG) is the number of range gates processed, λ is the wavelengthof the transmitted signal and ΔX is the map resolution size.Substituting into the foregoing expression, it may be seen that fortotal of 256 range gates a wavelength of 0.1 ft. and a resolution of 25ft., the total range slip during a dwell is approximately one-half of acell.

The detection of fixed targets depends on the capability of the radarsystem 10 (FIG.1) to make sufficient quality high resolution maps andthen, by signal processing techniques, to highlight potential targets onthese maps. It should be noted here that a 12 m. (40 ft.) resolutionarea search map will have sufficient image quality (even in a severerain environment) to make both natural and man-made features such asroad intersections, mountains and rivers clearly visible on the display.A 6 m. resolution map, on the other hand, is needed to ensurerecognition of targets such as bridges, hangars and water tanks. Inhigher resolution map a bridge contrasts markedly from background (madeup of usually of water, railroad tracks or ravines over which thebridges pass); hangars are detectable for like reason because of theirproximity to runways and aprons; and water tanks are readily detectablebecause they generally produce a regular geometrical high intensity mapimage.

Stationary vehicles, however, present a different type of problembecause the dimensions of such vehicles are usually in the same orderas, or smaller than, the dimensions of a cell when a 6 or a 12 meterresolution is used. Although an ultra-high resolution map, say one witha resolution of 1 to 2 meters, could be generated to allow vehicles tobe recognized, such a map would be too detailed to be used by anoperator in a tactical situation. That is to say, there would be so manyobjects displayed, each of which could be a vehicle by probably is not,that an operator would take too much to identify a vehicle, or vehicles.

The herein-contemplated radar system 10 (FIG. 1) employs a screeningprocess to winnow out, from a large number of displayed objects in anultra-high resolution map, those objects which most probably arevehicles. The screening process then, obviously, aids the operator byreducing the number of objects to be recognized and classified. Thescreening process will be described in detail hereinbelow. Suffice it tosay here that the contemplated screening process subjects all theprocessed radar return signals (which have been converted to digitalnumbers representing magnitude in a range-Doppler matrix) to a sequenceof logical tests including: (a) an amplitude threshold test, (b) a localmaxima test, (c) a clear region test, and (d) a scintillation test. Thelogical tests are such that only radar return signals which are probablyfrom vehicles will pass predetermined set limits. Before testing apreliminary process is carried on to determine that the digital numbersbeing tested are not derived from jamming signals.

Referring now to FIG. 11, a block diagram representation of the signalprocessing flow in the contemplated fixed target search mode ispresented. As may be seen, the raw (unprocessed) radar return datacorresponding to each of the four transmitted carrier frequencies fromeach of the antenna quadrants 14A, 14B, 14C and 14D, (FIG. 3A), as wellas the data from the sidelobe channel (not numbered) are stored inmemory (not numbered). It should be noted here that the data stored inmemory (not numbered) have been previously motion compensated by beingcomplex multiplied by a series of phase rotation multipliers as, forexample, the REPLICA lock data. The first step in the contemplatedsignal processing procedure is to form the monopulse sum (Σ) signal foreach of the four transmitted frequencies from the antenna quadrant data.The resulting Σ channel data for each of the four frequencies areseparately processed by means of an FFT algorithm to produce fourseparate range/Doppler matrices (maps). The resulting maps are theprocessed through two different signal processing paths, both of whichinvolved an editing process.

The first or ECM editing process examines each of the four maps to seeif any portions thereof are obscured or obliterated from effects of ECMas, for example, by means of active repeater jammers and/or spot noisejammers. Each of the maps is also examined on a cell by cell basisagainst data from the sidelobe monitor (not numbered) which is providedin the sidelobe channel and which has been separately FFT processed. Thesidelobe monitor data editing will be described in detail hereinbelow.Suffice it to say here that the sidelobe monitor editing is effective toreduce the sensitivity of the radar system 10 (FIG. 1) to the effects ofrepeater type jammers.

After ECM editing the data corresponding to the four separate maps areadaptively compressed from twelve bit magnitude words to four bitmagnitude words in order that they may be displayed in an optimum manneron the video display unit (not shown). The adaptive compression processinvolves rearranging the data samples in each of the four maps inaccordance with their amplitude distribution. That is, the samples arerearranged in a new matrix wherein amplitude is the abscissa and thenumber of samples is the ordinate. The amplitude corresponding to thegreatest number of samples is then used to form the mean and four bitmagnitude words formed about such mean. The resulting four bit magnitudemaps are then incoherently added to form a single display map. Theincoherent addition tends to fill in the gaps in the individual mapsidentified in the ECM editing process.

The second or fixed target detection editing process submits the data ineach of the four maps to: (a) an amplitude threshold test, (b) a localmaxima test, (c) a clear region test, and (d) a scintillation test. Theamplitude threshold test removes most elements within the range-Dopplermatrix from further consideration. The surviving members are thensubjected to a local maxima test wherein it is required that the elementof the map under consideration be greater than its immediate neighborssince it is desirable that selected target stand out from its adjacentneighbors to ensure that a point target is not counted more than once.

Each of the elements that survive the local maxima test are thesubjected to clear region test which requires that the local maximum begreater by fixed ratio than each of the targets falling in a ring aroundthe given target. This screening test favors point targets over spreadtargets and also tends to select the largest target in a givenneighborhood.

Those elements which survive the clear region test are then subjected toa scintillation test by comparing the four separate maps. Thescintillation test requires that the three immediately preceding maps beinvestigated to determine the presence or absence of a target which haspassed the clear region test. The question to be determined is whetheror not the target under test it also present in one, two or three of thepreceding maps at an amplitude which is within a given number of dB ofits magnitude in the fourth map. In the scintillation test lower andupper threshold are computed which are above and below the magnitude ofthe target under test. This maximum is then tested to determine if itlies within the adaptive bounds. The location of the maxima furnishesthe center point around which the next preceding map is tested. Thisprocess is iterated for three maps preceding the map under test. Itshould be noted here that all the cells tested in the scintillation testare in the same range bin as the target under test and that to pass thescintillation test the target must fall within the threshold on threeout of four maps.

An acceleration drift test is also employed to examine the Doppler driftbetween time sequential source maps. Interpolated Doppler frequenciesfor the cells surviving the prior tests are determined for each of thetime sequential maps. The individual cell acceleration consistency istested and the cell drift rate is tested against the composite mean.Those cells that fluctuate excessively or that exhibit drift rates whichdeviate sufficiently from the average values will also be deleted.

The foregoing logic testing can be extended to include some additionalECM editing features, as, for example, discrimination against cornerreflector decoys. As is known, corner reflectors are utilized tosimulate large fixed targets in an attempt to prevent the detection ofactual targets. However, corner reflector decoys may be discriminated byvirtue of their small scintillation vis-a-vis actual tactical targetsas, for example, tanks or trucks. Further, the clear region ratiotesting criteria can be modified to detect extended or shaped targets.Geometric pattern shape groupings of cell-sized targets can also befurther tested in a correlation process to evaluate the grouping as apossible convoy or surface-to-air missile (SAM) site.

Each of the four maps are also edited with respect to the sidelobemonitor data to remove jammer-induced false targets as was describedabove. Following editing with the sidelobe monitor data, the four mapsare incoherently added to form a composite map. In order for thepotential target in the composite map to be automatically designated toan operator, that target must meet the following criteria in thecomposite map: (a) the magnitude of the eight adjacent cells must beless than or equal to the magnitude of the test cell; (b) at least fiveout of eight of the adjacent cells must be less than half the magnitudeof the test cell; and (c) the magnitude of the nine cells contiguous tothe cells adjacent to the test cell must be no greater than one-quarterthe magnitude of the test cell. Further, a test cell that passes theforegoing criteria must not have a magnitude in any of the four mapsmaking up the composite map which is less than half that of itsintegrated scaled value on the composite map. The target search in thecomposite map is initiated at the center of the map and proceeds to mapextremities using a spiral search pattern.

Targets passing the foregoing tests are cataloged by range, angle,Doppler and magnitude and are subsequently stored within the digitaldata processor 30 (FIG. 1). Such targets are also automaticallydisplayed to the operator who may cursor a particular target for displayon a higher resolution map. The cursor addresses the location of theselected target within the digital data processor 30 (FIG. 1).

Moving Target Detection (FIG. 11A)

The radar system 10 (FIG. 1) must also detect ground moving targets inan environment of extended ground clutter, large discrete scatterers,wind-driven rain and chaff and active jamming. In such an environment,targets with one percent relative velocity must be detectable in theforward (±20°) sector, while targets with radial velocities as low as 1m/s must be detectable in the squint (20° to 90°) sector. The lattercase involves the detection of moving targets which are competing withclutter scatterers well inside the radar antenna sum beam.

Referring now to FIG. 11A, the contemplated moving target detectiontechnique will be described. Before proceeding, however, it will beinstructive to note that the effects of ground clutter and discretescatterer interference may be appreciated by recognizing that coherentpulse Doppler radars provide high resolution synthetic aperture mappingby virtue of the fact that each fixed ground point scatterer thatreflects energy has a Doppler phase history corresponding to its meanangular position relative to the aircraft velocity vector. Movingtargets impart an additional Doppler phase history to the radar signal;thus, the Doppler history from a moving target does not correspond tothe moving target's angular position with respect to the velocityvector, V. In many cases there will exist a ground clutter scattererwhich has a Doppler history due only to the motion of the aircraft 302(FIG. 10) which essentially corresponds with the Doppler history of amoving target. The angle, as viewed from the aircraft, between themoving target and such a ground clutter scatterer is referred to as thedisplacement angle. The displacement angle is, as previously explained,attributable to the Doppler ambiguities resulting from the radialvelocity of the moving target and the aircraft line-of-sight crossvelocity in the slant plane, and may be expressed as:

    Δα=V.sub.T /V.sub.AC sin α               Eq. (15)

Where V_(T) is the radial velocity of the target and V_(AC) sin α is thecross velocity of the aircraft 302 (FIG. 10). If a displacement angle ofsignificant magnitude exists, antenna weighting can be used to attenuatethe competing clutter scatterers such that the moving target becomesdetectable.

If the displacement angle is large relative to the transmitted antennabeamwidth, then conventional Σ channel detection techniques can beemployed. That is, the competing clutter cell will be attenuatedrelative to the target by virtue of the fact that it is on the far mainbeam skirts or in the sidelobe region when the target is within the mainlobe. Sum channel detection methods also apply when the total Dopplershift of the ground moving target is beyond foldover (i.e., greater thanthe maximum Doppler clutter frequency). Strictly speaking, there is nocompeting ground clutter when this occurs although either noisemodulated clutter residues or ambiguously sampled clutter may bedetected along with wind-driven rain and chaff. Thus, when required,rain rejection is provided by means of an adaptive polarizationtechnique which was described hereinabove.

Before proceeding, it should be noted here that during a given radardwell the antenna 14 (FIG. 1) will be slued to maintain theline-of-sight vector on a single area target focal point. As the antenna14 (FIG. 1) is slued, the PRF of the radar system 10 (FIG.1) must bechanged accordingly. It should also be recalled that during a givenradar dwell the PRI of the latter is controlled, for motion compensationpurposes, such that equal angle increments are swept out between pulsetransmission times. Finally, is should be noted that for long rangeapplications a relatively low radar PRF is utilized which may result inDoppler ambiguities in the case of high speed targets. From theforegoing it should now be appreciated by those of skill in the art thatthe contemplated moving target detection technique must be adaptive inorder to enable the radar system 10 (FIG.1) to detect both low and highspeed targets in a relatively complex clutter background.

Illustrated in FIG. 11A is a block diagram representative of thecontemplated signal processing technique for the ground moving targetdetection mode. Similar to the fixed target detection mode, the rawtarget return data (from each of the four antenna quadrantscorresponding to each of the four transmitted frequencies) are complexmultiplied by a series of phase rotation multipliers as, for example,the REPLICA signal, and the phase shifts required as a result of slueingor rotating the antenna 14 (FIG.1) during the radar dwell, as well asother motion compensation phase multipliers, and are ultimately storedin memory.

Following the phase correction multipliers the data are passed to anadaptive clutter canceller wherein a displaced phase center cluttercancelling technique is utilized to cancel background clutter. At thispoint it will be instructive to refer back for a moment to FIG. 3A wherethe arrangement of the antenna quadrants is illustrated. The phasecenters of antenna quadrants 14A and 14B are disposed along thehorizonal axis (not numbered) of the antenna 14, while the phase centersof antenna quadrants 14C and 14D are disposed along the vertical axis(also not numbered) of the antenna 14. The phase centers of antennaquadrants 14C and 14D must be phase compensated to account for theirdifference in elevation vis-a-vis the horizontal axis (not numbered) ofthe antenna 14. One-half the sum of the two latter is designated "E".

Referring briefly now to FIG. 7A, the contemplated clutter cancellingtechnique for the case of a two phase center antenna will be explained.It should be recalled here that the following technique was described indetail hereinabove with reference to the motion compensation sectionwhere the technique was expanded to include a three phase centerantenna. Suffice it to say here that if the phase centers are originallyat A_(o) and B_(o), then according to FIG. 7A the best time to take theB₁ sample for subtraction from A_(o) is when B crosses R_(Ao). The PRIof the radar system 10 (FIG. 1) is controlled such that equal angleincrements are swept out by the antenna 14 (FIG. 1) between pulsetransmission times. This ensures that when the second pulse istransmitted at time t₁, the B phase center will be on the line betweenthe phase center A at time t_(o) and the reference point, o; i.e. ,phase center B is somewhat along R_(Ao). The exact distance, L₁, thatphase center B at the time t₁ is from phase center A at time t_(o) mustbe known so that the phase of the samples obtained from phase center Bat time t₁ can be shifted by 4π L₁ /λ in order to match those obtainedfrom phase center A at time t_(o). The requisite phase shifts areobtained by generating the range, R_(Bo) (t), between the B phase centerand the map reference point, O, from the antenna velocity, radialacceleration and squint angle and sampling it each time a pulse istransmitted.

Once the radar return data has been properly phase compensated, thecharacterization of clutter rejection is dependent, inter alia, on thephysical separation of the phase centers being cancelled. As a generalrule, the greater the separation between the phase centers beingcancelled, the greater the degree of slow target detection. It will alsobe shown in detail hereinbelow that multiple antenna phase centers maybe utilized to distinguish between fast and slow moving targets.

If the returns from phase center A are to be cancelled with the returnsfrom phase center B, the procedure involves cancelling (subtracting) apresent sample from phase center A with a previous sample obtained fromphase center B. The particular phase center B sample chosen forcancellation (meaning the sample that corresponds to a previouslytransmitted pulse) is dependent on the distance between the A and Bphase centers and here is the second previously transmitted pulse. Thus,samples obtained from phase center A at time t_(n) are cancelled withphase center E samples obtained at time t_(n-1). In like manner, ifphase center E samples are to be cancelled with phase center B samples,then phase center E samples obtained at time t_(n) are cancelled withphase center B samples obtained at one time t_(n-1).

Digressing here now for a moment, it should be recalled that each fixedground point scatterer that reflects energy has a Doppler phase historycorresponding to its mean angular position relative to the velocityvector of the aircraft 302 (FIG. 10), and that moving targets impart anadditional Doppler phase history to the radar signal. The relative angleas viewed from the aircraft 302 (FIG. 10) between a moving target and aground scatterer having an identical Doppler frequency as the movingtarget is known as the displacement angle. As mentioned hereinbefore, ifa sufficient displacement angle exists, antenna weighting can be used toattenuate the competing clutter scatterers such that the moving targetbecomes detectable.

Referring now to FIG. 13, the effectiveness of the contemplated antennaweighting technique in detecting both fast and slow moving groundtargets will be explained. Thus, the solid curve of FIG. 13 representsthe difference pattern obtained by subtracting the returns from phasecenter B from those received by phase center A. The ordinate of FIG. 13represents received signal strength, while the abscissa representsDoppler frequency or, equivalently, angular extent from a given nullposition. The returns from any ground clutter scatterer located in thenulls will be completely cancelled. As explained hereinabove, any groundmoving target will impart an additional Doppler frequency to the radarsignal and, therefore, the Doppler frequency from a moving target willnot correspond to its angular position with respect to the null axis.Thus, if the total Doppler frequency of a moving target is sufficient toplace the target return in the Doppler filter corresponding to the nullaxis, that target will be detected. This is the so-called moving targetanomaly detection technique which is described in co-pending patentapplication Ser. No. 582,965 entitled "Moving Target Indicator (MTI)Radar System", filed May 30, 1975, inventors Hiller et al., now U.S.Pat. No. 4,217,583 issued Aug. 8, 1980. It is important to note herethat although the anomaly detection technique permits the detection ofmoving targets, further processing is required to accurately positionthe target in angle. It should also be noted that if a moving target isactually located in the reference null or is displaced in angle from thereference null axis or Doppler filter by an amount equal to the width ofthe difference pattern lobe, (i.e., it falls in the second differencepattern null), it will not be detected. It can be shown that thatangular displacement here corresponds to the displacement angle.Furthermore, as the displacement angle is related to the relativevelocity of the target, antenna weighting can be utilized to distinguishbetween various speed ground targets. The spacing between the nulls ofthe A-B pattern is determined by dividing the operating wavelength, λ,by the distance between phase centers A and B, which is here 12λ. As thedisplacement angle is equal to the difference between the nulls, it canbe seen that ground moving targets having a velocity of 8 percentrelative to the cross velocity of the aircraft 302 (FIG. 10) are likelynot to be detected by means of the A-B pattern. Thus, ground movingtargets having a velocity of up to 8 percent of the aircraft crossvelocity will be detectable by means of the A-B pattern. Thiscorresponds to the so-called "slow moving target" detection mode.

If, on the other hand, the returns from phase center E (formed bycombining the returns from phase centers C and D in a manner describedin detail hereinabove) are subtracted from those of phase center A toform a new null axis, the difference pattern indicated by the dashedline of FIG. 13 is obtained. As may be seen, the difference pattern lobeof the A-E pattern is twice as wide as that of the A-B pattern. Thisfollows from the fact that the distance between the A and E phasecenters is 6λ. Ground moving targets having a relative velocity of up to16 percent of the cross velocity of the aircraft 302 (FIG. 10) willtherefore be detectable with A-E pattern. That is to say, the angularextent or Doppler frequency excursion between the nulls of the A-Edifference pattern corresponds to a target having a velocity equal to 16percent of the cross velocity of the aircraft 302 (FIG. 10). This is theso-called "fast moving target" detection mode. It should again be notedhere that the three phase center clutter cancellation techniques justdescribed simply allow for the detection of either slow or fast movingground targets, and does not provide for the location in angle of thedetected targets. Additional processing, to be described in detailhereinbelow, is required to accurately position the detected targets inangle.

Referring back now to FIG. 11A, following the adaptive cluttercancelling phase processing described hereinabove, the datacorresponding to each of the four transmitted frequencies for each ofthe antenna phase centers A, B and E are processed by an FFT algorithm,the net effect of which is to arrange the data in a range/Dopplermatrix. Following the FFT processing, the data undergo a thresholdingprocess wherein the difference patterns A-B, A-E and E-B are formed.Those returns that exceed a predetermined threshold level in the A-E andE-B patterns are classified as fast moving target candidates, whilethose exceeding a similar threshold in the A-B pattern are classified asslow moving target candidates. The moving target candidates identifiedby the thresholding process are next subjected to a series of editingprocesses designed to filter out false targets induced by ground basedECM equipment.

The first of the editing processes is a so-called "SKIRT EDITOR"designed to identify and remove those false targets introduced into theradar system 10 (FIG. 1) by repeater type jammers. As is known, thelatter attempt to introduce false targets into the radar system 10(FIG. 1) via the sidelobes of the antenna 14 (FIG. 1). If the returnsfrom the repeater jammers occur at frequencies corresponding to theDoppler frequency of the returns from the main lobe of the antenna 14(FIG. 1) or at frequencies corresponding to the Doppler ambiguities ofthe main lobe, then the false returns could be mistaken for realtargets. The skirt editing process is designed to examine the sidelobesof the various antenna patterns formed to test for the presence of falsetarget returns. Thus, the returns as seen by a single antenna quadrant,here quadrant A, are compared with the returns as seen by the sum ofantenna quadrants A and B, (A+B), as well as the sum of all four antennaquadrants, (A+B+C+D). It will be appreciated by those of skill in theart that the pattern from the single antenna quadrant A is sufficientlybroad to cover the null in the A+B pattern, which results from thephysical arrangement of antenna quadrants A and B, and the sideloberegion of the A+B+C+D pattern. In the contemplated skirt editing processan arbitrary threshold is established based on the magnitude of thereturn signals as seen by the single antenna quadrant A. That thresholdis here two times the magnitude of the returns seen by antenna quadrantA. Hence, if twice the magnitude of the returns seen by antenna quadrantA in any given region is greater than the magnitude of the returns seenby either the A+B or A+B+C+D patterns in the same region, then thereturns from the latter (sum) patterns not exceeding the 2A thresholdare rejected as false targets.

It should be noted here that the just described skirt editing processmay be expanded, depending on the threshold level selected, to providesome degree of protection against main lobe repeaters or break-lockrepeaters which initially repeat the radar signal faithfully to capturethe tracking circuits of the radar system 10 (FIG. 1) and then move offthe true target return, either in range or frequency, thereby pullingthe radar system 10 (FIG. 1) off the true target return signal.

The next step in the editing process is also an ECM editor referred toas the sidelobe monitor. The sidelobe monitor editing is similar to theskirt editing just described in that both are designed to reject thereturns from repeater type jammers which attempt to enter the radarsystem 10 (FIG. 1) via the antenna sidelobes. It should be noted herethat the intent of these repeater type jammers is to produce multiplefalse targets in the radar system 10 (FIG. 1) in an attempt to exceedthe data handling capability of the latter and thereby prevent truetargets from being identified. In the skirt editing process justdescribed the returns detected in a single antenna quadrant pattern werecompared against the returns seen by the patterns resulting from the sumof antenna quadrants A and B as well as antenna quadrants A, B, C and D.The sidelobe monitor editing provides not only protection againstrepeater type jammers from entering the sidelobes of the single quadrantantenna pattern, but also a degree of protection against spot noisejammers. Thus, it will be recalled that the sidelobe monitor (notnumbered) described hereinabove with reference to FIGS. 4A and 4B has anadaptive null forming capability which may be used to cancel the returnsfrom spot jammers.

It should be noted here that the sidelobe monitor data are receivedthrough the auxiliary channel of the RF receiver 18 (FIG. 1) and areseparately FFT processed into a range-Doppler matrix. The resulting map(range-Doppler matrix) is compared against similar maps generated fromthe data received by each of the antenna quadrants. The sidelobe monitormap is then compared with each of the quadrant maps and any potentialtarget whose amplitude in the former is greater than in any of thelatter is rejected as a false target. This editing is dependent on thefact that the gain of the sidelobe monitor pattern is greater than thatof the antenna quadrant patterns in the sidelobe region of the latter.

Following the sidelobe monitor editing the surviving target candidatesare subjected to an additional ECM editing stage designed to nullify theeffects of swept, frequency modulated (FM) barrage type jammers on theradar system 10 (FIG. 1). As is known, the effect of this type of jammeron the latter is to reduce the signal-to-noise ratio to an extentdetermined by the effective radiated power of the jammer, the range tothe jammer, and the gain of the antenna sidelobes in the direction ofthe jammer. One known method to counter the effect of such a jammer isto turn off (blank) the R.F. receiver 18 (FIG. 1) whenever the jammersweeps through the radar band. An alternate technique which takes fulladvantage of the adaptive null forming capability of the hereincontemplated radar system 10 (FIG. 1) to discriminate against a numberof spatially separated slow sweeping barrage jammers would be to form aspatial null in the direction of each jammer as they randomly sequencethrough the radar spectrum. This technique presupposes the existence ofa means of cataloging the location of the slow sweeping jammers whichhere may be provided by means of the sidelobe cancelling circuits (notnumbered) described hereinabove with reference to FIGS. 4A and 4B. Thecontemplated null forming technique will be described in detailhereinbelow with reference to the target angle estimation section.Suffice it to say here that that technique takes advantage of themultiple antenna phase centers to form a pair of steerable monopulsenulls.

If, however, blanking must be utilized for some reason as, for example,the sheer number of swept jammers that must be nullified, then theherein contemplated radar system 10 (FIG. 1) still provides anadvantage. That is to say, systematic blanking disturbs the continuityof the received waveform which will result in the introduction ofDoppler sidelobes in the return signal. However, since the blankingcommands are developed within the radar system 10 (FIG. 1), thesignature (time of occurrence and duration) of the blanks may be storedwithin the latter and upon completion of the radar dwell time thisblanking sequence (signature) may be transformed into weights and usedto reduce the generated Doppler sidelobes to an acceptable level.

The potential moving target candidates that survive the foregoing ECMediting processes are next subjected to an angle gate thresholdingprocess, sometimes referred to as a spread moving target gate, which isdesigned to remove those potential moving target candidates which areattributable to intrinsic clutter. The effect of intrinsic clutter whichmay, for example, be due to the motion of wind driven trees in leaf, isto cause an increase of energy in a particular Doppler filter or aspread of energy between a group of Doppler filters. The angle gatethresholding technique therefore subjects each moving target candidate,which will be recalled is positioned in a range Doppler matrix, to amonopulse angle measurement. That measurement takes full advantage ofthe multiple phase center antenna in forming the monopulse angleestimate. Thus, the estimate is formed by normalizing the difference ofthe E-A and B-E difference patterns to the sum of those differencepatterns and may be expressed as:

    ((E-A)-(B-E))/((E-A)+(B-E))                                Eq. (16)

The procedure now is to form the foregoing ratio for every candidatetarget in the range Doppler matrix. As is known, normalization willassign each moving target candidate in the range/Doppler matrix (map) toa particular Doppler filter or angular location. If two or more targetcandidates are assigned to the same Doppler filter or angular location,then the angle gate editor will reject all of these candidates as beingintrinsic clutter due, for example, to the motion of a single tree. Inlike manner, if some number n of target candidates, where n>4, areassigned to adjacent Doppler filters or angular locations, the anglegate editor will reject all of these candidates as being intrinsicclutter due, for example, to the motion of a group of wind driven trees.

Before proceeding, it should be recalled here that the radar system 10(FIG. 1) is required not only to detect ground moving targets, but alsoto accurately position those targets on a diaplayed map. Thus, thosemoving target candidates that survive all of the foregoing editingprocesses are subjected to a so-called "three phase center" monopulsetarget detection technique. As previously explained, the radar returndata may be combined to provide a minimum of three colinear phasecenters A, B and E which lie in the slant plane determined by theaircraft velocity vector and the line of sight vector to a groundreference point. In the three phase center monopulse target detectiontechnique the returns from the three phase centers are processed suchthat the angles to both a target and its competing clutter patch can beresolved. The solution is directly analogous to multiple sidelopejammers in which more than one jammer is cancelled in the sidelobes of atracking radar by using multiple small apertures, one for each jammer.The three phase centers form a pair of independently controllabledifference patterns exhibiting two nulls, one for the target and one forclutter. Error signals can then be obtained for either or both thetarget or clutter by using the appropriate source in the denominator ofthe three phase center monopulse normalization routine.

As the moving target candidates that survive the foregoing editingprocesses are arranged in a range-Doppler matrix (map), an estimate ofthe Doppler frequency associated with each target cell is readilyobtained. With knowledge of the Doppler frequency associated with thepotential target cell, the angular location of that cell with respect tothe antenna boresight axis may be determined by means of the followingexpression:

    f.sub.D =2V sin BΔB/λ                         Eq. (17)

where f_(D) is the Doppler frequency associated with the target cellrelative to boresight, V is the aircraft velocity, B is the antennaaspect angle with respect to the aircraft velocity vector, λ is thewavelength of the transmitted signal and ΔB is the angular location ofthe target cell.

The first step in the three phase center target detection process is toplace the nulls obtained from both the A-E and E-B patterns at theangular location of the target cell. It should now be appreciated bythose of skill in the art that the contemplated null steering may beaccomplished by shifting (rotating in angle) the data from each of theantenna phase centers A, B and E prior to forming the difference patternbeams. The requisite amount of phase rotation is given by:

    φ=2πd sin ΔB/λ                         Eq. (18)

where d is the distance between the antenna phase centers and ΔB is theangular location of the target cell. It should also now be appreciatedthat when both difference pattern nulls are placed on the given targetcell (Doppler filter) any clutter within that cell (Doppler filter)will, for all practical purposes, be completely nulled. Therefore, anyresidual energy within that cell (Doppler filter) will be due to themotion of an off axis moving target. The angular position of the movingtarget relative to the null axis may be estimated by means of aconventional monopulse (Δ/Σ) measurement made on the reference cell(Doppler filter). That is to say, once the monopulse discriminator curveis formed the off axis location of the target may be estimated from themagnitude of the residual energy within the target or reference cell.The procedure now is to phase shift (rotate in angle) one of thedifference pattern nulls by an amount given by:

    φ.sub.T =2πd sin ΔB.sub.T /λ           Eq. (19)

where ΔB_(T) is the estimated off axis angular location of the movingtarget. Next, with one of the difference pattern nulls located on thecell (Doppler filter) corresponding to the estimated position of themoving target, the reference cell (Doppler filter) is re-examined. Ifthe residual energy within the latter falls to the level of the null oris otherwise indiscernible, then the moving target is unambiguouslylocated in angle. That is to say, if the second difference pattern nullis accurately positioned on the moving target, then the latter will beunable to contribute any energy to the reference cell or Doppler filter.If, on the other hand, when the second difference pattern null is placedon the estimated moving target cell there is little or no change in theenergy level within the reference cell (Doppler filter), then the movingtarget is not accurately located and the process is repeated.

Once the correct angular location of the moving target is determinedthat location is stored in memory (not shown) for subsequent display.The three phase center monopulse detection technique is then repeatedfor each of the moving target candidates that survived the editingprocesses. When all the moving target candidates have been accuratelylocated in angle a composite, incoherent map is formed from the returnscorresponding to each of the transmitted frequencies. The composite mapis displayed to an operator who may then select a particular target orgroup of targets for attack.

PPI Mode (FIG. 1)

The radar system 10 (FIG. 1) is required to provide a ground mappingmode utilized principally for course navigation and in the designationof sectors for high resolution mapping. In such a mode, sometimeshereinafter referred to as the PPI mode, a noncoherent frequency agilesector map is generated. The requisite maps must cover a 60° azimuthsector and must be displayed in any one of three operator-designatedrange scales (i.e. 15 to 100 km, 5 to 40 km, or 5 to 20 km). The rangeresolution varies with the range scales being respectively 150, 60 and30 m. It will be appreciated by those of skill in the art that in thelong range (15 to 100 km) application the standard antenna Σ beam willbe sufficiently broadened (due simply to antenna beam dispersion withrange) to provide the requisite elevation coverage. In consequence,then, in the long range PPI mode the antenna 14 need only be stabilizedin elevation while it is slued in azimuth to provide the requisiteterrain coverage.

In the short range (5 to 20 km) PPI mode a different situation arises.That is, the Σ beam will not be sufficiently dispersed with range toprovide the requisite elevation coverage. In this instance, advantage istaken of one of the antenna segments, here segment 14AS, to form a fanbeam. As is known by those of skill in the art, a fan beam will beformed from an antenna where one of the principal dimensions of suchantenna exceeds the other by a ratio in the range of from 2 to about 5.The length of the antenna segments 14AS, 14BS, 14CS and 14DS exceedstheir width by the required amount and, therefore, each of such segmentswill form a fan beam. Thus, in the short range PPI mode the antenna 14(FIG. 1) is rotated to align antenna segment 14AS in a vertical positionand the four axis gimbal assembly 12 is used to stabilize the antenna 14at a constant elevation angle irrespective of aircraft motion. Elevationmonopulse calculations are then performed on selected range bins andthese monopulse estimates, as a function of range, are employed toestimate and track the height of the aircraft 302 (FIG. 10) above themapped terrain. This information is employed in controlling the antennaelevation fan beam pointing. It should be noted here that, as mentionedbriefly hereinabove, the nominal PRF of the radar 10 will be varied as acosine of the antenna pointing angle to compensate for the change inrange with antenna scan angle.

Antenna Segment Null Patterns (FIGS. 3A, 4A, 4B)

It should be recalled here that the antenna aperture is subdivided intofour identical quadrants 14A, 14B, 14C and 14D and four identicalsegments 14AS, 14BS, 14CS and 14DS. Further, as mentioned hereinabove,the antenna aperture (not numbered) is divided into square quadrantsrather than serrated quadrants even though serrated quadrants wouldprovide low quadrant sidelobes in the 0°, 45° and 90° planes of theantenna 14. As also noted, if the quadrants 14A, 14B, 14C and 14D wereserrated the grating lobes would be randomly distributed to either sideof the principal and diagonal planes of the antenna 14. The squarequadrant was here chosen because, in such a configuration, the sidelobesare known to be along the diagonal planes and, therefore, compensationfor such sidelobes may be effected.

Referring now to FIG. 4B, it should be recalled that the output signalsfrom antenna segments 14AS, 14BS are passed to switch networks 245V,245H and that such switch networks may, in response to a control signalprovided by the radar synchronizer 34 (FIG. 1), produce an output signalcorresponding to: (a) the input signal received from antenna segment14AS; (b) the input signal received from antenna segment 14BS; or (c)the difference between the input signals received from antenna segments14AS and 14BS. The output signals from the switch networks 245V, 245Hare ultimately passed through the auxiliary channel (not shown) of theR.F. receiver 18 (FIG. 1) to the signal processor 26 (FIG. 1) and thedigital data processor 30 (FIG. 1).

It should now be appreciated by those of skill in the art that the mainlobes (fan beam) from antenna segments 14AS, 14BS will be of greatermagnitude vis-a-vis the antenna quadrant sidelobes that reside in theirrespective diagonal planes. In consequence, then, the returns fromantenna segments 14AS, 14BS may be utilized to cancel jamming signalswhich would otherwise enter the radar system 10 via such diagonal planesidelobes.

It should also be appreciated that the pattern resulting from taking thedifference of antenna segments 14AS, 14BS will have a null located atthe geometric center of the antenna 14 (corresponding to the location ofthe Σ beam peak) with lobes extending radially outward in each of thediagonal planes. Such a pattern is effective in cancelling jammers thatwould normally fall beyond the range of the quadrant sidelobecancellers. The segment difference pattern offers a second advantage inthat it provides jammer cancellation without increasing the clutterlevel as the clutter is along the range axis.

Difference Channel Sidelobe Cancelling

Referring back now for a moment to FIGS. 4A and 4B, it should berecalled that the arithmetic networks 259V, 259H are included in thesidelobe cancelling channels (not numbered) to form both the monopulsesum and difference signals from the quadrant sidelobe cancellingelements (not numbered). It should also be recalled that switches 261V,261H are provided to selectively apply either the monopulse sum ordifference signals through the R.F. receiver 18 (FIG. 1) to the digitaldata processor 30 (FIG. 1) for further processing. The reason forselectively applying either the monopulse sum or difference signals fromthe sidelobe cancelling elements to the digital data processor 30(FIG. 1) will be appreciated when it is recognized that conventionalsidelobe cancellers utilize only a single beam to cancel the returnsfrom the sidelobes of both the sum and difference beams of the antennathat they are intended to protect. Thus, while such single beam sidelobecancellers are effective in cancelling undesired returns from thesidelobes of the main antenna sum beam, they introduce a tracking errorwhen they are utilized to cancel undesired returns from the main antennadifference beam sidelobes. This tracking error results from the factthat the beam from such conventional sidelobe cancellers is designed tobe of greater magnitude than the sum beam sidelobes in the region of thesum beam sidelobes. In consequence, then, the peak of the sidelobecanceller pattern will be coincident with the null axis of the mainantenna difference beam and, therefore, interference signals from thesidelobe canceller beam will be injected into the main antennadifference channels. The radar system will then cause its differencechannel null to be shifted in an attempt to null these error signals,thereby introducing tracking errors into the antenna system.

The problem in the herein contemplated radar system (wherein the returnsfrom each of the antenna quadrants 14A, 14B, 14C and 14D (FIG. 3A) aremodified by their sidelobe cancellers before being combined to formmonopulse sum and difference beams) is analogous to the foregoingproblem. Thus, while each of the quadrant cancellers is controlled toprovide the optimum degree of cancellation for each of the antennaquadrants, the phase and amplitude settings of each of the quadrantsidelobe cancellers will not be identical due principally to the actualphysical separation between each of the quadrant sidelobe cancellers. Asa result, then, when the returns from, say, antenna quadrants 14A and14B are subtracted to form the monopulse azimuth difference beam it ispossible for the sidelobe cancellation components in each of thechannels to combine in such a manner as to introduce an interferencesignal in the difference channel. In light of the foregoing, themonopulse arithmetic networks 259V, 259H are provided in the sidelobecancelling channels (not numbered) so that a monopulse differencepattern null may be formed from each of the quadrant sidelobe cancellersand steered in the direction of the interference. When the quadrantdifference pattern is then formed, the effect of the sidelobe cancellersin the resulting difference channel will be minimized.

Ultrahigh Resolution Mapping (FIG. 14)

The herein-contemplated system is required to generate a high resolution(2 ft.) map around an area designated by the radar operator. This modeis utilized for target classification whereby vehicles may bedistinguished from each other. A chirp waveform must be used to achievethe wide bandwidth required to achieve the desired degree of resolution,along with equalization of the waveform time sidelobes using the replicalock equalization technique which was mentioned before hereinabove.Briefly, that technique involves taking a sample of the transmittedwaveform and using that sample to correct errors in the returned pulse,thereby to reduce the time sidelobes of the compressed pulse and tocorrect random pulse-to-pulse phase errors.

It will be appreciated by those of skill in the art that the focus zoneof the generated map would be an unacceptably small area around thecenter of focus if the map of aircraft motion effects are not properlycompensated. Thus, the techniques of equiangular sampling, which wasdescribed hereinbefore, and polar format interpolation, which will bedescribed in detail hereinbelow, are used to expand the focus zone intoa usable size. The polar format technique inherently implies the use ofa two-dimensioned FFT for processing since corrections must be made inthe complex plane before resolving the map into individual cells. Inorder to be able to process the two dimensional map data in a reasonableamount of time, the technique of A/D strobe frequency control, which wasdescribed briefly hereinabove, is used to align the data samples takenin such a way that they can be processed with a minimum amount ofinterpolation and therefore in a minimum amount of time.

Even with the compensations which are made for aircraft motion, errorscan cause areas of the map to be improperly focused when there are largeaccelerations. Thus, a technique of self-focusing, wherein point targetsdetected in the map are used to determine the extent of the accelerationerror so that the error may be removed before the final FFT processwhich resolves all the cells, is used to correct for accelerationerrors.

Digressing here now for a moment, it should be recalled that the replicalock technique is used to solve two problems produced by distortion ofthe wideband chirp signal in the high power microwave portion of theradar system 10. This distortion takes the form of amplitude and phasemodulations which are impressed on the transmitted waveform producingdeviations from the ideal quadratic phase characteristic of a chirpsignal. These deviations, if not compensated for in some manner,generate time (range) sidelobes and Doppler sidelobes which degrade thesmall area contrast ratio of the map. The second problem which is solvedby the replica lock technique is the removal of pulse-to-pulse randomphase changes. The average phase of the replica is measured for eachtransmitted pulse and the deviation from nominal phase is removed in thesignal processor 26 (FIG. 1). The replica-lock technique takes samplesof each transmitted pulse for the purpose of measuring the average phaseof the pulse. For the purpose of compensating for the waveformtransmission characteristics, the signal processing necessary to derivethe correction factor is only done once per dwell as the phasetransmission characteristics of the transmitter and receiver componentsare slowly changing functions.

When the replica is used to compensate the received signal for time(range) sidelobes, the compensation consists of multiplying thedigitized radar data by the reciprocal of a stored replica of thetransmitted pulse. Since both quantities are complex, this operation toa first order is a series of phase rotations which correct for phaseirregularities in the transmitted signal.

As is known, defocusing of the map is caused by quadratic phase terms inthe recorded data for points displaced from the map center which remainafter the focus compensation for the center point has been performed.While equiangular sampling does, in fact, linearize the phase historyfor points displaced from the map center, its effectiveness is limitedin very high resolution applications by differential range shifts ofpoints within the map during the mapping dwell. The differential rangeslip problem may be understood by considering that as the syntheticaperture map is developed the radar range gate is fixed on a centralpoint. Points at the same range but different azimuth positions havevarying ranges with respect to this point and, over the dwell, they willmove into and out of the central gate. If a simple range gated system isused these targets will smear through several range gates, and adefocused map will result. The relative target motion at differentazimuth angles is the parameter used to obtain azimuth resolution and itis implicit in all synthetic aperture maps. For ultra high resolutionmaps the differential motion of λ/2 (where λ is the wavelength) requiredto resolve two adjacent cells is about 1/40 of the cell size and sinceseveral hundred azimuth cells are desired, a differential range slip ofseveral cells for points at the edges of the map is inherent andunavoidable.

Polar formating is a signal processing technique used to solve thedifferential range slip problem. The concept of polar formating evolvedout of optical processing where initially the dechirped, uncompressedreturns for each transmitted pulse are recorded directly on film and thereturns from successive pulses are laid down in parallel strips. Theresult is a hologram of the ground image. Coherent processing of thehologram (equivalent to a two-dimensional FFT) recovered the groundimage. When ultra high resolution was attempted it was found that layingthe strips down with a slight angle between successive strips (insteadof parallel) would compensate for differential range slip and give acorrect hologram of the ground image. This slight angle betweensuccessive strips was equivalent to using a polar coordinate system forrecording and became known as "polar formating". The herein contemplatedsystem uses a discrete digital equivalent of an optical polar formatingtechnique which has been optimized for the real time operationalrequirements of the system. The technique employs the following sequenceof operations. First, a long frequency ramped signal is transmitted andthe received signal is heterodyned against a similarly ramped localoscillation. In digitizing the return data the timing of the A/Dconverter strobe pulses is varied by means of the frequency slippedclock signal developed within the synchronizer 34 (FIG. 1) to compensatefor changes in the line of sight depression angle during the dwell. ThePRI being varied, data samples equally spaced in angle around the mapcenter are produced. The digitized return data are subsequently thinnedin a presummer/roughing filter to lower the sample rate. Next,interpolation of the data is performed to produce a modified data set inwhich the nonlinearity of the phase history of any point in the map isminimized. Finally, a two-dimensional FFT is performed on the modifieddata set to generate the ultra high resolution maps.

It should be noted here in passing that prior to interpolation thereturn data consists of a fan-shaped grid of points in which the dataarranged in vertical columns are separated by a constant angle as theresult of PRI control and the data arranged in horizontal rows areseparated by a constant distance as the result of A/D strobe control.Interpolation across horizontal points is then used to effectivelyarrange the data in a horizontal grid.

The effect of the foregoing sequence of operations is to generate asampled data phase plane across which each resolvable point in the mapproduces a unique, nearly linear phase gradient. Since the effect ofinterpolation is to rearrange the data into the form of an orthogonalgrid, a two-dimensional FFT performed along the two principal axes willproduce a map in which the two dimensions are also orthogonal. That is,the cross-coupling between the range and the cross-range dimensionsproduced by differential range slip is eliminated by the interpolationprocess.

As previously mentioned, the times at which A/D strobe pulses and radartransmitter pulses are produced are both controlled to simplify thecomputations involved in producing polar formatted data. The equationwhich governs the timing of the A/D strobe pulses is: ##EQU8## wheref_(o) =transmitter frequency

K=chirp slope

C=velocity of light

Δ=map resolution

θ=line-of-sight azimuth angle to the map reference point

φ=line-of-sight elevation angle to the map reference point

r_(y) =range to the map reference point

The stobe timing consists of a constant term and a linear term. Theconstant term gives the time delay to the first data sample taken afterany transmitter pulse, while the linear term gives the repetitioninterval of the strobe pulse train. Since both θ and φ can be consideredconstant during any radar interpulse interval, the strobe pulsefrequency is constant during any receive interval. The dependence of tupon cos φ is such as to maximize the extent of the focused region inthe presence of vertical maneuvers.

Before proceeding with a detailed description of the contemplatedinterpolation process, it will be instructive at this point to recallthat the result of stretch processing of the chirp waveform from a pointtarget located within the mapped region is a relatively long fixedfrequency tone whose frequency is a function of the range to the target.The received tones are sampled with an A/D at a fixed rate with theresult that the point target is thereby converted to a phase gradient inthe sampled domain for that pulse. This process is repeated over manypulses and a two-dimensional sampled phase plane results.

Since the PRF is significantly higher than the frequency extent of theultra high resolution (UHR) map, the initial returns are over sampled inthe azimuth direction. Consequently, these data are thinned in thepresummer/roughing filter to obtain a lower sample rate. In order tosuppress aliasing around the filter output sample lines, the roughingfilter is sampled at approximately twice the bandwidth occupied by themapped region which restricts the phase quadrant (the phase differencebetween successive data points) to ±90° for points at the edges of themap.

The basic equation for the phase, P, of a given sample is: ##EQU9##where: N=number of samples in the range dimension

N_(y) =y position of point target expressed as an output index number

n=input index number in range

M=number of samples in the azimuth dimension

M_(x) =x position of point target expressed as an output index number

m=input index number in azimuth

B=resolution factor (resolution÷λ/2)

If the middle term in the foregoing equation were zero, the phase wouldbe the linear superposition of two independent functions and twoorthogonal FFTs could be performed to obtain the map. The center term isnot zero except, of course, for M_(x) =0 at the map center. This is dueto differential range slip and is, as explained hereinabove, inherent inall synthetic aperture maps.

The function P_(n),m can be converted to a function P_(n),k of the form##EQU10## Since, strictly speaking, the radar system 10 (FIG. 1)operates on complex voltages, the index, k, may be treated as anoninteger with different gradient values depending on n. A fractionalcounter (not shown) may then be used to increment k. Samples forinterpolation may then be selected by the integer portion of thefractional counter and the weights to be applied by the fractional partof the counter. For a given n, the interpolation process will generateharmonics. The nature of the harmonics, which have frequencies that area function of n, is that when the range FFT is performed, they tend tosmear into several Doppler filters. This smearing tends to reduce theharmonic power in a single Doppler filter.

As mentioned briefly hereinabove, the contemplated UHR signal processingmode utilizes a self-focus technique to correct for quadratic phaseerrors in the map data introduced by inertial instrument drift. Ifuncompensated, these errors would degrade the focus quality of the mapin the cross-range dimension thereby increasing the amplitude of thenear-in sidelobes and ultimately broadening the cell width.

The self-focus technique consists of dividing the data record obtainedfrom a mapping dwell into subrecords in which the quadratic phase erroris small enough to be ignored. Each of these subrecords is subsequentlyused to generate a map in which strong point targets can be identified.Although the resolution of these maps will be poorer than that specifiedfor the UHR mode, it will still be sufficient to permit an accuratedetermination of target location in each of them. The principal effectof the quadratic phase error produced by instrument drift is to causethe images of fixed point targets to change their locationssystematically in successive maps. Thus, it is possible to estimate themagnitude of the error by tracking the apparent motion of these targets.The contemplated self-focus technique extracts acceleration correctioninformation in this manner and applies these corrections to the map datagathered during the entire dwell to form a well focused, high resolutionmap from the corrected data.

Having described a preferred embodiment of the contemplated system, itwill now be apparent to one of skill in the art that many changes may bemade without departing from out inventive concepts. That is to say, theactual structural elements making up a system according to our inventiveconcepts may be changed from those illustrated so long as the resultingsystem is operative as described. In particular, the provisions heremade for: (a) compensating for the error due to the presence of a radomeand for vibration of the antenna; (b) the assembly, in effect, of anumber of radars using a single array antenna assembly with a"four-axis" gimbal to make up a monopulse radar which may be operated asa synthetic aperture radar on a maneuvering aircraft; (c) the"three-phase" monopulse arrangement to improve operation in the presenceof clutter; (d) the arrangement of a sidelobe cancelling antenna locatedat the phase center of each quadrant of the single array antennaassembly to render the system almost impervious to electroniccountermeasures which would incapacitate any known system; and (e) themethods used to generate and to process signals so that maps withvarious degrees of resolution may be generated even in the presence ofjamming signals, broadly distinguish our inventive concepts. It is felt,therefore, that this invention should not be restricted to its disclosedembodiment, but rather should be limited only by the spirit and scope ofthe appended claims.

What is claimed is:
 1. The method of calibrating nominally identicalreceiver channels in an airborne radar using a phased array antennadivided into quadrants to permit monopulse operation when desired, suchantenna being gimballed within a radome, such method comprising thesteps of:(a) periodically orienting the phased array antenna to directthe boresight line of such antenna toward a predetermined point on theradome; (b) isotropically radiating, from the geometric center of thephased array antenna, a pilot pulse to provide, at the phase center ofeach quadrant of such antenna, an equi-phase and equi-amplitude signalat the operating frquency of the airborne radar; (c) processing theequi-phase and equi-amplitude signal at each phase center in a differentone of the receiver channels; and (d) comparing the output signals fromeach one of the receiver channels to a corresponding reference signal toderive the desired calibration signals in accordance with thedifferences existing between each one of such output signals and thecorresponding reference signal.